Self-interference cancellation antenna systems and methods

ABSTRACT

The present application describes systems and methods of performing self-interference cancellation. Such systems may include generating a transmit signal along a transmit path of a transceiver, where the transmit signal can be sent through a circulator to isolate the transmit signal from a receiver. The transmit signal may be transmitted from an antenna, and a signal may be reflected from the antenna, where the reflected signal may be at less power than an incident power to the antenna, and where the reflected signal may include a transmitter carrier signal and a transmitter noise. A received signal may be routed from the antenna to the receiver, the reflected signal may be routed through a filter and a phase shifter, and the signal may be combined with the received signal in the receive path to cancel the portion of the transmit signal that entered the receive path towards the receiver from the circulator.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application claims priority to U.S. Provisional Application No.62/193,148, filed on Jul. 16, 2015, the disclosure of which isincorporated herein by reference in its entirety.

FIELD

The present invention generally relates to radio frequency systems, inparticular to self-interference cancellation antenna methods andsystems.

BACKGROUND

In high power radio applications, a very large and expensive filter isoften used to separate transmit and receive signals. This is especiallytrue when a system consists of a high power transmitter as well as asensitive receiver where the transmit and receive bands are very neareach other.

In a typical radio, there is a transmit chain and a receive chain. Thetransmit chain will have an associated gain, which will directly amplifythe thermal noise. This thermal noise can fall in the receive band andmust be filtered. In radio systems that have complex modulatedwaveforms, intermodulation noise from the power amplifier can fall intothe receive band in addition to the amplified thermal noise of thetransmit chain. In other cases where a more efficient class of amplifieris used in conjunction with pre-distortion, the pre-distorter can addnoise as well. These noise sources place a requirement on the filter forthe amount of rejection required in the receive band to block thetransmitter from degrading the sensitivity of the receiver.

Another issue present in a typical radio system is that of the transmitcarrier power getting into the receiver. This power is not in band andthus can be filtered out, but it must be filtered enough that none ofthe receiver chain saturates. This places an additional limitation onthe filter.

Accordingly, there is a need for improved self-interference cancellationantenna methods and systems. Various aspects of the disclosure may solveone or more of these problems and/or disadvantages.

SUMMARY

In one aspect, the disclosure describes a self-interference cancellationsystem, including a transmitter power amplifier configured to amplify atransmit signal. A circulator may be coupled to the transmitter poweramplifier, where the circulator may be coupled to atransmission-reception path of the self-interference cancellationsystem. In some embodiments, an antenna may be at an end of thetransmission-reception path, where the antenna is configured to transmitthe transmit signal and receive a receive signal. A receiver amplifiermay be coupled to a reception path of the self-interference cancellationsystem, and a phase shifter may be coupled to the reception path.

In another aspect, the disclosure describes a method of performingself-interference cancellation including the steps of generating atransmit signal along a transmit path of a transceiver, and sending thetransmit signal through a circulator to substantially isolate thetransmit signal from a receiver, where at least a portion of thetransmit signal enters a receive path towards the receiver. A signal maybe reflected from the antenna, where the reflected signal is atsubstantially less power than an incident power to the antenna, andincludes a transmitter carrier signal and a transmitter noise. Incertain embodiments, a received signal may be routed from the antenna tothe receiver, and the reflected signal may be routed through a phaseshifter in the receive path, where the reflected and phase shiftedtransmitter noise may be combined with the received signal in thereceive path to cancel the portion of the transmit signal that enteredthe receive path towards the receiver from the circulator.

It is contemplated that in certain aspects the disclosure describes aself-interference cancellation system, including a transmitter poweramplifier configured to amplify a transmit signal, a first circulatorcoupled to the transmitter power amplifier, the first circulator beingcoupled to a transmission-reception path of the self-interferencecancellation system, and an antenna at an end of thetransmission-reception path, wherein the antenna is configured totransmit the transmit signal and receive a receive signal. In someembodiments, a receiver amplifier may be coupled to a reception path ofthe self-interference cancellation system. A second circulator may becoupled to the first circulator, and a filter may be coupled to thesecond circulator in the reception path.

In yet another aspect, the disclosure describes a method of performingself-interference cancellation including the steps of generating atransmit signal along a transmit path of a transceiver, and sending thetransmit signal through a circulator to substantially isolate thetransmit signal from a receiver, where at least a portion of thetransmit signal enters a receive path towards the receiver. A reflectedsignal may be generated from the antenna, where the reflected signalfrom the antenna is at substantially less power than an incident powerto the antenna, and the reflected signal may include a transmittercarrier signal and a transmitter noise. In certain embodiments, areceived signal may be routed from the antenna to the receiver, and thereflected signal may be routed through a filter, where a reflectedtransmitter carrier signal is passed through the filter to a load. Thereflected transmitter noise may be combined with the received signal inthe receive path to cancel the portion of the transmit signal thatentered the receive path towards the receiver from the circulator.

There has thus been outlined, rather broadly, certain embodiments of theinvention in order that the detailed description thereof may be betterunderstood, and in order that the present contribution to the art may bebetter appreciated.

BRIEF DESCRIPTION OF THE DRAWINGS

In order to facilitate a more robust understanding of the application,reference is now made to the accompanying drawings, in which likeelements are referenced with like numerals. These drawings should not beconstrued to limit the application and are intended only to beillustrative.

FIG. 1A illustrates an example of interfering energy.

FIG. 1B illustrates the topology according to an aspect of thedisclosure.

FIG. 1C illustrates signal data associated with removal of carrier powerby the carrier cancellation loop.

FIG. 1D illustrates signal data associated with the removal ofself-interference from the receive band.

FIG. 1E illustrates another exemplary topology and elements of feedforward cancellation.

FIG. 2 illustrates receiver based processing, according to an aspect ofthis disclosure.

FIG. 3 illustrates a tone generation function used as part of thedigital residual interference cancellation loop according to an aspectof this disclosure.

FIG. 4 illustrates positioning of tones overlaid on the receiver signalaccording to an aspect of this disclosure.

FIG. 5A illustrates a digital interference cancellation system accordingto aspect of this disclosure.

FIG. 5B illustrates an exemplary method that accounts for variations inthe received signal according to an aspect of the disclosure.

FIG. 5C illustrates a self-interference channel estimation method inaccordance with an aspect of this disclosure.

FIG. 5D illustrates the operation of the complex multiplier inaccordance with an aspect of this disclosure.

FIG. 5E illustrates a method for in-phase and quadrature phase (IQ)compensation in a full duplex transceiver in accordance with an aspectof this disclosure.

FIG. 6 illustrates architecture to invert distortion, in accordance withan aspect of this disclosure.

FIG. 7 illustrates architecture reducing an error signal using feedbackloops, according to another aspect of this disclosure.

FIGS. 8A and 8B illustrate measured data with respect to both magnitudeand phase across the receive band, in accordance with an aspect of thisdisclosure.

FIGS. 9A-D illustrate the degree of improvement in isolation obtainablefor different combinations of amplitude error and/or phase error, inaccordance with an aspect of this disclosure.

FIG. 10 illustrates data where carrier cancellation has occurred usingfeed-forward self-interference cancellation but without application ofthe digital residual interference cancellation, in accordance with anaspect of this disclosure.

FIG. 11 illustrates operation with both the carrier cancellation anddigital interference cancellation loops applied, in accordance with anaspect of this disclosure.

FIG. 12A illustrates a self-interference cancellation system accordingto an aspect of the disclosure.

FIG. 12B illustrates a self-interference cancellation system accordingto another aspect of the disclosure.

FIG. 13A illustrates diagram of a self-interference cancellation circuitwith two circulators, a phase shifter, and a filter according to anaspect of the disclosure.

FIG. 13B illustrates a method of self-interference cancellationaccording to an aspect of the disclosure.

FIG. 14 illustrates diagram of a self-interference cancellation circuitwith two circulators, a phase shifter, and a filter according to anotheraspect of the disclosure.

FIG. 15 illustrates diagram of a self-interference cancellation circuitwith two circulators and a filter according to an aspect of thedisclosure.

FIG. 16 illustrates diagram of a self-interference cancellation circuitwith one circulators and a phase shifter according to an aspect of thedisclosure.

FIG. 17 is a diagram illustrating a frequency hopping method, accordingto one aspect of this disclosure.

FIG. 18 is a flowchart illustrating a method of operation of theapparatus of this disclosure, according to one aspect of thisdisclosure.

FIG. 19 is a flowchart illustrating a method of operation of theapparatus of this disclosure, according to one aspect of thisdisclosure.

FIG. 20 is a flowchart illustrating a method of operation of theapparatus, according to one aspect of this disclosure.

FIG. 21 is a flowchart illustrating a method of operation of theapparatus, according to one aspect of this disclosure.

DETAILED DESCRIPTION OF THE ILLUSTRATIVE EMBODIMENTS

A detailed description of the illustrative embodiments will be discussedin reference to various figures, embodiments and aspects herein.Although this description provides detailed examples of possibleimplementations, it should be understood that the details are intendedto be examples and thus do not limit the scope of the application.

Reference in this specification to “one embodiment,” “an embodiment,”“one or more embodiments,” “an aspect” or the like means that aparticular feature, structure, or characteristic described in connectionwith the embodiment is included in at least one embodiment of thedisclosure. Moreover, the term “embodiment” in various places in thespecification is not necessarily referring to the same embodiment. Thatis, various features are described which may be exhibited by someembodiments and not by the other.

Generally, many cellular and other communications systems operate in afull duplex manner—simultaneous transmission and reception—usingdifferent frequency bands to transmit and receive, known as FrequencyDivision Duplex (FDD). FDD refers to the physically-definedfrequency-based division between uplink and downlink, as opposed to timedivision, code division, polarization division, spatial division, andrelated variants. A frequency band is the set of allowed operatingfrequencies for communications. As illustrated in FIG. 1A there is afrequency separation between transmit and receive bands. In a wirelesstransceiver, information is transmitted at a power level that istypically many times higher than the received power. Interfering energy,which is generated by the transmitter nonlinearities and other noiseprocesses, is leaked into the receive band frequencies where itinterferes with desired reception. This interference is referred to asself-interference in this description. Proper receiver operationrequires attenuation or transmit-to-receive isolation of thisself-interference, in many cases on the order of 100 dB or more.

In one aspect, a RF front end solution for a FDD wireless system isprovided. This solution reduces some of the entire isolation requirementon the duplexer and can eliminate the duplexer in many cases. This isaccomplished by use of interference cancellation. The methodincorporates multiple cancellation methods which may beseparately-employable methods: 1) the receiver, and 2) DigitalCancellation which digitally samples and subtracts the interference fromthe desired signal at the receiver. Feed-forward Cancellation andDigital Cancellation can be applied independently in given applicationsor the two methods may be combined. Also antenna techniques may beapplied in a given application. By stacking the isolation obtainablewith the multiple cancellation methods and antenna techniques, isolationlevels in excess of 100 dB are obtainable.

In an embodiment, the first method is based on the use of analog meansfor generation of an inverted signal which approximates the transmitsignal received in the receive band and which is then subtracted fromthe transmit signal before it reaches the antenna and receiver. Thispart of the schema accommodates cancellation of signals at high powerand with higher interference levels. The resultant signal afterfeed-forward cancellation contains residual energy from the transmitsignal which may then be removed using another method such as antennaisolation or the second Digital Cancellation method.

In another embodiment, the digital self-interference means includes achannel estimation schema which properly determines the magnitude andphase response across the bandwidth of the channel and uses thisinformation to model and generate an Infinite Impulse response, IIR,filter to generate an interference (e.g., interference) cancellationsignal which is subtracted from the residual signal after Feed-forwardCancellation to obtain the signal of interest received from sourcesother than the transmit signal of the base station or wirelesstransceiver.

These methods may be used in tandem with each other and may be used withdifferent antenna methods to achieve high levels of isolation and enableoperation at high power levels over wide bandwidths.

The following objectives are met by the subject matter of thisapplication:

1. Ability to cancel enough TX-RX interference (aka self-interference)to eliminate filters in systems with additional isolation methods suchas TX-RX antenna separation, digital cancellation, or receivercancellation.

2. Ability to cancel enough self-interference to reduce the requirementson filters used in traditional one-antenna systems. Therefore, thefilters can be made smaller and cheaper.

3. Utility for existing FDD systems (like 3G and 4G cellular) and unlikeexisting methods can be applied to other systems for which waveforms areyet to be standardized, potentially with a mere software update.

4. Utility of the method to allow some systems to be designed withoutfilters or other band-specific components, enabling band-independenttransceivers or software defined RF front ends.

5. Novel transceivers that operate at many frequencies, or evenuser-defined frequencies, with an insignificant increase in size evenwhen combined with software defined radios

6. Novel secure radios that operate over a great number of frequenciesor bands even when combined with frequency-hopping software definedradios.

7. When combined with LTE-Advanced systems, enables band aggregationand/or frequency hopping in support of the LTE-A standard.

8. For low power systems (picocell, femtocell, cellular handset), thesubsystem can be microfabricated and integrated on chip or bysystem-on-chip methods.

9. For high power systems (microcell, macrocell), the subsystem can beconstructed with off-the-shelf components as part of an RF circuit

10. Advantageous leverage to more capable devices.

11. Use one or a combination of methods to enable fullystandards-compliant wireless base stations for use in environments whereoutside interferers are low, e.g., high-altitude wireless base stations.

General Architecture

The power amplifier (PA) generates a high-power transmit (downlink)waveform, but its inherent nonlinearity generates intermodulation of thetransmitted energy. For example, when a cellular base station isconsidered, the transmission is at the downlink frequency. For userequipment, such as for example a mobile phone, the transmitter would beon the uplink frequency. Some of this energy is present within thereceive frequency band and, unless filtered using for example a duplexeror other filtering arrangements in present art systems, will interferewith the desired receive signal. Several factors determine the amount ofinterference present in the receive band, including but not limited to:

Duplex spacing. The closer in frequency the RX band is to the TX band,the higher the intermodulation interference.

Amplifier topology. A linear (class A or AB) produces the leastinterference but consumes the most DC power. Higher power consumptionincreases heat sink size. Doherty PAs are efficient but narrowband andnot as linear.

Amplifier device technology. Commercial LDMOS devices produce the leastinterference but are narrowband. Gallium Nitride (GaN) devices can helpmake broadband PAs but are less linear, resulting in more interference.

Modulation bandwidth. Larger data rates have created the requirement tosupport larger bandwidths, providing more spectral energy to fall intothe receive band. Single-carrier GSM has most of its energy containedwithin 200 kHz, UMTS a little less than 5 MHz, and LTE has increasedchannel bandwidth allocation up to 10 MHz, 15 MHz, and 20 MHz, dependingon RF band.

Feed Forward Self-Interference Cancellation

In an aspect of the application, a feedforward method forself-interference cancellation may direct signals through multiple paths(e.g., loops) to reduce interference, such as for example, interferenceas result of a power amplifier. In one example, a first path implementscarrier cancellation to eliminate high power transmitter carrier outputfrom a copy of captured power amplifier interference. This first pathmay be designated as a carrier cancellation path. A second path adjustsphase and amplitude of a waveform before injecting it into a mainforward path. This second path may be designated the error cancellationpath (also discussed herein as interference cancellation path) thatcompensates for the phase delay and amplitude of the residual signal inthe receive band after the transmit carrier is cancelled. When the firstpath and the second path are combined the result may be a high-powertransmitter that is effectively ultra linear in the receive band, asinterference from the transmitter in the receive band is removed.Isolation between transmit and receive bands on the order of 30 dB to 40dB may be obtained via the use of feedforward RF cancellation.

FIG. 1B illustrates an exemplary topology and elements of feed forwardcancellation. Although discussed in more detail herein, below is asummary of the paths for FIG. 1B. In FIG. 1B, the carrier loop iscomposed of the path that contains the sampled carrier energy, startingat coupler 102, adjusted in amplitude and phase by LVM 111, amplified bypower amplifier 112, optionally filtered by filter 113, and subtractedfrom the sampled carrier+noise power in segment 122. The forwardtransmission path consists of the main forward path through coupler 102,the bandpass filter 103, the transmit power amplifier 104, and thecoupler that samples carrier+interference (noise) (coupler 105). Thenoise or error cancellation path is the main forward path throughcoupler 105 and coupler 106 and the path of the sampled interferencethrough 105 (samples carrier plus interference), coupler 119 where thecarrier energy is subtracted to leave essentially only the sampledinterference, LVM 116 which adjusts magnitude and phase of the sampledinterference, amplifier(s) 117, and coupler 106 where sampled andmodified interference is subtracted (added in equal magnitude and 180°out of phase) from the energy in the main forward path 120.

At 100, TX signal 101 which corresponds to the waveform applicable for agiven air interface, such as WCDMA, HSPA, or LTE, generated upstream ofthe feed-forward cancellation is input to the feed-forward cancellationelectronics of FIG. 1B. Filter 103, which may be a fixed filter in abank of filters or a tunable filter, or omitted, may filter the transmitband (TX signal 101) before being amplified by power amplifier 104.Power amplifier 104 amplifies the filtered TX signal 101. The output ofpower amplifier 104 is a higher power signal (e.g., signal 108) whichincludes interference that may be generated by power amplifier 104. Theinterference may result from the inherent non-linear nature of poweramplification done by power amplifier 104.

In addition, coupler 102 samples TX signal 101. TX signal 101 has itsamplitude and phase changed (e.g., using a vector modulator—linear VM111—as well as power amplifier 12) so that the magnitude of the sampledsignal as injected by coupler 119 (e.g., signal 114) is the samemagnitude as signal 115 and the phase offset as discussed in more detailherein. One purpose for filter 103 is to help provide matching of groupdelay in the main forward path when compared to the carrier loop, iffilter 113 is used. Filter 113, for example, may bandpass-filter usingtuned or a bank of filters, or omitted, resulting in signal 114. Signal114 has the same magnitude as signal 115, but is 180 degrees out ofphase after any phase shift from coupler 119. For example, if thecoupler is a balanced Wilkinson, then signal 114 and signal 115 have thesame magnitude and are 180 degrees out of phase. On the other hand, ifcoupler 119 is a 90° hybrid, then signal 114 and signal 115 have thesame magnitude and are 90 degrees out of phase (the path design would besetup so that the extra 90° for signal 114 causes signal 114 and signal115 to be 180° out of phase when they combine). Signal 115 is obtainedvia coupler 105 which samples signal 108 which includes power amplifiergenerated interference.

Injecting of signal 114 with signal 115 on segment 122 provides for theremoval of the transmit energy of signal 115. For additionalperspective, loop 98 retrieves a sample of the undistorted transmitterin order to provide a signal that removes the transmit energy from thesample of the noise in loop 99. Without the loop 98, loop 99 may addeven more noise via power amplifier 117 associated with the existence ofthe transmit energy. The sample of the transmit energy in loop 98 isadjusted in amplitude and phase such that the transmitter energy removedat coupler 119. Without the transmit energy as described, loop 99 maytake a more faithful sample of the interference, with minimal additionof its own distortion and with acceptable linearity specifications forthe components in loop 99.

The signal with the removed transmit energy on segment 122 is input tolinear vector modulator (LVM) 116 and amplified by power amplifier 117.The output of power amplifier 117 is signal 107 for which the transmitsignal has been removed, but for which the interference in the receiveband that was generated by the power amplifier remains. Signal 107 is anappropriately phase-shifted and amplified signal that provides for thecancellation of sample noised in the receive band when injected viacoupler 106 into the main path 120. This signal 9 is a high powerversion of TX signal 101 with interference cancelled in the receive bandas the transmit carrier has been cancelled in the receive band.

Moreover, noise that remains in the receive band may be cancelled bydigital residual interference cancellation as discussed in more detailherein (e.g., FIGS. 5A-5E discussed below). In summary, an input signalfrom tone generation functions used for channel characterization,automatic gain control, and I/Q compensation in the digital method maybe injected via a coupler onto segment 118. The resultant signal ofsegment 118 would then be coupled into the signal at the output of thepower amplifier on main path 120 via coupler 6 to obtain signal 109, thehigh power transmit signal which is fed to downstream to transmitantenna functions.

FIG. 1C illustrates an exemplary measured result of the remaining signalas seen in the receive band after carrier cancellation. The carrierpower is removed. Here, for example, trace 123 is the carrier signal inthe receive band that occurs in signal 108 and trace 124 is the signalin the receive band that occurs in signal 109.

FIG. 1D illustrates exemplary results measured across a wide bandwidthafter interference cancellation. FIG. 1C illustrates carriercancellation of over 23 dB, while the FIG. 1D illustrates interferencecancellation in the receive band. The noise from the power amplifier(e.g., power amplifier 104) is canceled by 30 dB. Cancellation isachieved over broad bandwidth, suitable for 4G waveforms and otherfuture broadband waveforms. Due in part to group delay matching in allloops, feed-forward self-interference cancellation overcomes limitationsof many conventional systems which operate over much narrowerbandwidths. In addition, feed-forward self-interference cancellation asdiscussed herein enables additional downstream processing using digitalprocessing means to deal with elimination of the residual noise aftercarrier cancellation where the residual noise occurs at low power levelscloser to the noise floor of the system.

Variations due to temperature and component tolerances should beaccounted for in implementations of feed-forward self-interferencecancellation. It may beneficial to use components that have especiallyhigh linearity, in order avoid limiting the cancellation achievable,especially for high power systems. The electronics of the feed-forwardself-interference cancellation path may drift with time and temperature,so an electronic feedback control circuit, such as controller 121, maybe used. Controller 121 may maintain carrier cancellation lock. Theresult is a sampled output of the interference from the Power Amplifieras seen in the receive band which remains stable throughout theoperation of the system. Implementations of the present invention haveconfirmed the operation of the art taught herein. It is beneficial forgroup delay to be carefully controlled for both the carrier cancellationand error cancellation loops. The carrier cancellation loop may driftsuch that the sampled carrier power may be out of exact anti-phase withthe sampled carrier+interference signal. Or it may have drifted to nolonger be the same amplitude as the sampled carrier+interference signal.In either case, carrier power will be less than optimally removed fromthe sampled interference output of coupler 119. The controller 121circuit senses the resulting carrier power (the power after subtraction)and adjusts amplitude and phase (e.g., in LVM 111) to maintain optimalphase and amplitude matching of the carrier loop.

Discussed below is one example for control of the carrier cancellationpath of the feed-forward self-interference cancellation path. First, thephase and magnitude of the LVM 111 in the carrier cancellation path areswept over their usable range to find an operating point that is closeto the optimal operating point of the system (e.g., the operating pointthat minimizes the carrier power at the output of the feed-forwardself-interference cancellation path). With regard to LVM 111, the powermeasurement for control is a broadband power measurement of the signalat the output of coupler 119. Once the sweep has completed, controller121 utilizes a “perturb and observe” algorithm to find the optimaloperating point. It changes the magnitude and phase by small steps; ifthe cancelled carrier power decreases by changing either value in agiven direction, then that direction is closer to the optimal operatingpoint. Otherwise, if the power increases, then the operating point is inthe other direction, and controller 121 switches directions of magnitudeand phase values. This may be done periodically.

Control of the error cancellation path using LVM 116, for example, issimilar to the carrier cancellation path, with the exception that powermeasurement of the error cancellation path is a sample of the outputsignal 109. This sample may be fed through a super-heterodyne receiverto limit the power measurement to the desired receive channel.Otherwise, control for the two methods is similar.

Controller 121 has control over the tunable, controllable or switchableelements shown in FIG. 1B and include magnitude and phase of LVM 111,LVM 116, filter 103, filter 113 (e.g., tunable or switched), or othercontrollable elements to be included as determined to be needed on anapplication-to-application basis. For example, filter 103 and filter 113shown in FIG. 1B may be used, but filter 103 and filter 113 are notnecessary in many applications. In addition, controller 121 may controlother elements such as FFTs, IIR filters, and other digital elementswhich discussed generally herein.

FIG. 1E illustrates another exemplary topology and elements of feedforward cancellation that is similar to what is shown in FIG. 1B. Someof the common elements as shown in FIG. 1B have the same numbers asshown in FIG. 1E. Many of the elements shown in FIG. 1E are optional andmay be removed. Elements (e.g., gains, delays, or power detectors) shownin FIG. 1E may be removed or repositioned to accommodate different goalsof effectiveness of the system.

As shown by the arrows, there are multiple alternative filter locations.Loop 98 (carrier cancellation loop) and loop 99 (interference/errorcancellation loop) of FIG. 1E have different elements than what is shownin loop 98 and loop 99 of FIG. 1B. System 95 of FIG. 1E provides anexample of the use of the feed forward cancellation method with thedigital cancellation method. The connection and configuration of system95 to the feed forward cancellation method is similar to digitalresidual interference cancellation loop 1206 as connected withfeed-forward self-interference cancellation loop 1204 shown in FIG. 12B.

With continued reference to FIG. 1E, in the interference cancellationloop (loop 99), there may be a component called a power detector thatgoes to a controller. The controller is sensing the analog power, whichis primarily the carrier power that makes it through the carriercancellation loop (loop 98). Carrier cancellation occurs in the coupler119. The residual amount that is not successfully canceled may be fedinto the optional power detector. The power detector gives an outputvoltage that responds to the carrier power. Controller 121 may be usedto minimize the voltage at that location. Controller 121 makes use ofthe output of the power detector. Controller 121 may adjust the amountof attenuation (attenuator 94) and the amount of phase shift (phaseshifter 93), such that it minimizes the amount of detected carrierpower, in order to stabilize the carrier cancellation loop (loop 98). Ifthe controller is digital then there may be one controller. There alsomay be multiple separate controllers when the carrier cancelation isdone in a fully analog way. The use of separate controllers may befaster and could prove advantageous. For example, in securecommunications there may be a need to change frequencies quickly and afast stabilization loop may be preferred with an analog method

Those skilled in the art will certainly recognize that elements of thefeed-forward self-interference cancellation of FIG. 1B and FIG. 1E maybe configured for applications for use with different air interfaces orRadio Access Technologies (RAT) and for operation in different bands aswell as at different power levels. And gains, coupling losses and poweramplifier and other gain elements as well as feedback control parametersmay be applied appropriately to a given application of the art of thepresent invention. For example, the disclosed components of the systemsand methods may be electronically reprogrammable via controller 121 sooperation may happen over many bands.

Digital Self-Interference Residual Cancelation

FIG. 2 illustrates receiver based processing which enables cancellationof Power Amplifier generated noise which occurs in the receive bandacross the receive band. Signal A, the transmit signal is input toantenna means for transmission of signals to subscriber units or othertransceivers. Tone generator generates two fixed tones which areconstant during the operation of the system as well as swept tones whichare periodically applied during the operation of the system to enablecharacterization of the channel between the transmitter and thereceiver. Tone generator includes up conversion functions to enabletransmitting the tones coincident with the band of operation of the RXsignal.

Further to FIG. 2, the Transmit signal is sampled via coupler 2 andinput to filter 3, which reduces the power of the Tx carrier. The outputof filter 3 feeds into low noise amplifier 7 to generate the RX1 signal10 which is a composite signal used in further digital process as areference signal which contains the tones which were injected into thetransmit path as well as the residual noise generated by the poweramplifier which occurs in the receive band.

As also shown in FIG. 2 a receive signal F which is obtained fromreceive antenna functions includes the residual noise from the poweramplifier which occurs in the receive band after transmit carriercancellation as well as the tones which were injected into the transmitsignal and also includes the desired receive signal from which residualnoise is to be further cancelled. The receive signal F is amplified vialow noise amplifier 8 to obtain the RX2 signal 9 which is furtherprocessed to remove residival transmit noise.

According to another aspect of the disclosure, digital self-interferenceresidual cancellation samples the output of the power amplifier beforethe antenna and then subtracts this sample from the received signal fromthe antenna. To do so, the system injects tones into the receive channelto characterize the frequency response of the channel with respect toboth amplitude and phase across the channel. With this characterizationinformation, the sampled data can be equalized to be the same as thereceived signal and subtracted successfully. In summary, injected tonesare a reference that goes through the same distortion as theself-interference signal, which allows for the inversion of thedistortion of the self-interference and, therefore, broadbandcancellation of the interference from the received signal.

Isolation between transmit and receive bands on the order of 20 to 40 dBmay be obtained with digital self-interference residual cancellation,depending on the degree of match in both phase and amplitude between thecompensating signal and the residual transmit signal across the receiveband channel as seen at the input to the receiver.

FIG. 5A illustrates an exemplary digital residual noise cancellationsystem (or generally, an interference cancellation system) used as astage of Tx-Rx self-interference cancellation before the received signalis processed by the physical layer of the radio. The digital residualnoise cancellation system uses a sample of the transmit signal beforethe transmit antenna of a device (e.g., signal based on the sample fromcoupler 2 of FIG. 2). The sample of the transmit signal is referred toas Rx signal 149, which includes the injected tones, and may be used todetermine which part of the received signal (Rx signal 131) is transmitinterference. Once the transmit interference is determined it may beremoved from the Rx signal 131. Rx signal 131 includes the interferenceplus the tones plus the desired signal. As discussed herein, todetermine and remove the transmit interference, the system continuouslyinjects two tones (constant during the operation of the system) justoutside the receive channel and periodically sweeps a third tone throughthe channel.

It has been determined that the injected tones add only a very smalldegradation to the desired signal. The levels used for the injectedtones are controllable and advantageously enables transmission of thesetones at levels which are just large enough to allow them to be seen inthe receive band, but small enough that they add minor degradation tothe operation of the system. The level of the tones may be significant.If the tones are too large then they degrade the performance of asystem, because there is limited dynamic range in the receiver. Thetones should be loud (i.e., large) enough to meet a signal to noiseratio that give valid results for the IIR filter (e.g., IIR filter 146).The tones should be loud (i.e., large) enough to meet a signal to noiserequirements of the rest of the system while respecting the dynamicrange limitations of the system For example, there may be a receiverthat has a 60 dB dynamic range and a goal may be to try to cancel 30 dBof interference. In this scenario the power of the tones can be adjustedsuch that they are above the interference level but below the 60 dBlimit, giving up to a 30 dB range of valid power levels, depending onthe required SNR. Therefore tones should be between 20 dB and 30 dBabove the interference level. The tones should be large enough so thatthey are greater than the interference level, but less than the dynamicrange of the receiver. The digital residual noise cancellation systemdiscussed herein may operate at baseband.

With continued reference to FIG. 5A, a first stage in the system 130 maycompensate for imbalances in I/Q compensator 132 and I/Q compensator 148for RX signal 149 and Rx signal 131, respectively, used to compensatefor gain and phase imbalances that might have occurred during downconversion using one of the fixed tones mentioned above (e.g., the fixedtone at the lower edge of the reception band). If these imbalances areleft uncompensated, they lead to skew and rotation in the receivedconstellations, which reduces the amount of cancellation. Thecompensation algorithm uses one of the two fixed tones mentioned above(e.g., the fixed tone at the lower edge of the reception band) andminimizes a power of the image tone at the negative of the frequency ofthe image tone.

Regarding IQ Compensators 132 and 148 (‘I’ for in-phase and ‘Q’ forquadrature phase, as known to one of ordinary skill in the art), thedistortion from IQ imbalance can be characterized by a matrix E=[1, 0;−g sin(φ)), g cos(φ)], where g is a gain mismatch between the two armsof the IQ compensator 132 and/or the IQ compensator 148 (each having arespective IQ demodulator), and φ is a phase mismatch between the twoarms. In one aspect, a processor (not shown) in a controller 121 isconfigured to compute the inverse of E (i.e., a matrix E⁻¹) and applythe inverse matrix to the received digital data so that the originalnon-distorted constellation can be processed. The inverse of E istherefore E⁻¹=1/(g cos(φ)*[1, 0; g sin(φ)), 1]. Further, anon-transitory or tangible computer-readable medium (not shown) may beconnected to the processor. Such computer-readable medium may storecomputer executable instructions, which when executed by the processorcause the processor to carry out the various features andfunctionalities of the disclosure. Such non-transitory computer-readablemedium may include memory devices such as read-only memory (ROM), randomaccess memory (RAM), and the like, or combinations thereof.

In an embodiment, a structure as shown in subsystem 600 in FIG. 6 can beused to invert the distortion in the received signal. The subsystem 600is included inside each of the IQ compensators 132 and 148. In FIG. 6, acomplex input signal S_(in) at an input 167 (representing the receivedsignal at a receiver) is first separated into its real and imaginaryparts or components in a module 168, which are respectively multipliedby coefficients c1 stored in a module 169 and c2 stored in a module 170,respectively, by multiplier 171 and adder 172. The imaginary componentobtained from the module 168 is combined with the output of themultiplier 172 by adder 173. The signals at the inputs of a combiner 604are both real signals. Therefore, the output of the adder 173 is real.The signal at the output of the combiner 604 is a corrected signal. Inother words, if the input 167 receives S_(in), then an output of blockthe combiner 604 is E⁻¹*s.

In one aspect of this disclosure, one or more feedback loops 702 and 704to calculate the coefficients c1 and c2 in FIG. 6 may be deployed. Herethe subsystem 600 is described with respect to the feedback loops 702and 704 in a subsystem 700 in FIG. 7. A fixed tone is injected in atransmission signal and an image of the fixed tone is observed in thereceived signal. For example, if the fixed tone injected is at afrequency f_(Tone), the image tone is at −f_(Tone), the goal of thefeedback loops is to minimize the power of the image tone.

This is accomplished in the subsystem 700 by taking a Fast FourierTransform (FFT) 708 after buffering the incoming signal 602 at a buffer706 and multiplying at a multiplier 710, the values of FFT bins 713 and714 by constants 716 and 718 corresponding to the fixed tone and itsimage tone. In one aspect, the buffers are optional and alternativefunctional elements may be used by one of ordinary skill in the art inview of this disclosure. The result is that if the image tone'smagnitude is dropped to 0, the product in complex form at a module 712will also be zero. The imaginary part of the output of the module 712 isfed to a negative unity gain buffer 720 of the feedback loop 702 and thereal part of the output of the module 712 is fed to a unity gain buffer722. The resulting error signal at an output of the multiplier 710 issplit into its real and imaginary parts, which are fed throughintegrators 724 and 726 (which are rate translated by rate translators728 and 730, respectively, and which are preceded by the buffers 720 and722, respectively as well as an adder 732 which adds an offset 734) toforce the DC error to 0. This is similar in concept to how a Costas Looplocks to an incoming complex signal and separates out the I and Qcomponents. In one aspect, the buffer 722, the offset 734, and the addermay be optional and may be replaced by a single negative unity gainblock.

In one aspect of this disclosure, the next stage of the system 130 isconfigured to equalize the differences in magnitude and phase variationseen between the Rx signal 131 and the Rx signal 149. Because theinterference from the transmitter 100 must be removed over the entirebandwidth of the receive channel shown in the subsystem 130, it is notgood enough to simply match the magnitude and phase at a singlefrequency and then subtract the two signals. To accomplish thischaracterization, a tone is periodically swept through the receivechannel at a discrete number of frequencies, and the power of that toneat each port is measured and processed. This tone is referred to as aswept tone, and the discrete frequencies may be equally spaced in thereceive band.

Measurements of magnitude and phase of the down converted swept tone ateach of the discrete frequencies are made via an FFT 133. Further,measurements are made at each of k discrete frequencies of the swepttone to determine a measured H_(k where:)

H₀ is the measurement of H (e^(jω) ₀)

$\mspace{20mu} \begin{matrix}\text{?} \\\text{?} \\{\ldots \mspace{11mu},{{and}\mspace{14mu} \text{?}}}\end{matrix}$ ?indicates text missing or illegible when filed

H_(k) is the measurement of H(e^(jω) _(k)), where H denotes a transferfunction such that H(e^(jw)) is the frequency-domain representation ofthe difference between the path from the output of the transmitter 100to Rx ports RX1 and RX2 in FIG. 5A receiving the signals 131 and 149. IfX(e^(jw)) as the frequency-domain representation of the transmittedsignal. Then R1(e^(jw)) is the portion of the transmitted signal that isreceived at Rx port RX1 and R2(e^(jw)) is the portion of the transmittedsignal that is received at Rx2. It will be appreciated that RX1 is theport that receives a sample of the transmitted signal and RX2 is thenormal receive port for the system. Then H is (R2/X)/(R1/X)=R2/R1 wheree^(jw) has been dropped for clarity of this equation defining H.

Advantageously, the value of k is controllable by the controller 121 fora given instance of the system 130. The higher the value of k, thenumber of measurement points across the receive band, the more likely itis that the IIR filter 146 will be able to match the frequency responsewell. By way of example only and not by way of limitation,implementations of the system 130 have shown that values of k as smallas 47 yield cancellation results which can meet applicationsrequirements in many applications.

Within software in the controller 121, tables of values of H_(k) forboth magnitude and phase are determined for a given measurement cycle(e.g., a swept tone measurement cycle for that of the swept tone)normalized to an angular frequency ω_(k). Here, ω_(k)2πf_(k)/Fs wheref_(k) is the frequency of a given discrete tone and Fs is the samplingfrequency.

FIGS. 8A and 8B illustrate the measured data with respect to bothmagnitude and phase across the receive band. The plots define thechannel response estimate for the channel between the transmitter andthe receiver. In FIG. 8A, a best-fit curve shown as the continuous linecurve for the magnitude of the samples of the received signal (e.g., theRx signal 149). The samples are illustrated as circles in the curve inFIG. 8A. Likewise, in FIG. 8B, a best-fit curve shown as the continuousline curve for the phase of the samples of the received signal (e.g.,the Rx signal 149). The samples are illustrated as circles in the curvein FIG. 8B.

Further elements shown in FIG. 5A enable the generation of acancellation signal which is inverted and added at an adder 135 to theRx signal 131 to cancel the residual transmitter noise to obtain adesired signal 134 which is then provided to further components of thereceiver for demodulation and further baseband processing. An InfiniteImpulse Response (IIR) filter 146 generates an IIR response for each ofthe Rx signal 131 and the Rx signal 149 and periodically updates thisresponse during the operation of the system 130 in order to properlyfilter the interference seen in the Rx signal 149 with respect tomagnitude and phase across the reception band. Generation of thecoefficients of the IIR filter 146, which has p zeroes and q poles, pand q being integers, is done in conjunction with a least-squaresestimator 147 which attempts to provide the best fit of the IIR responseto the measured response of the system 130 over the receive band.

The Infinite Impulse Response Filter transfer function in the z-domainis represented by:

G(z)=(a ₀ +a ₁ z ⁻¹ + . . . +a _(p) z ^(−p))/(1+b ₁ z ⁻¹ + . . . +b _(q)z ^(−q))   (1)

The Equivalent IIR filter transfer function in the frequency domain isrepresented by:

G(e ^(jω))=(a ₀ +a ₁(cos ω−j sin ω)+ . . . +a _(p)(cos pω−j sin pω)/(1+b₁(cos ω−j sin ω+ . . . +b _(q)(cos q ω−j sin qω)   (2)

The IIR transfer filter can also be represented by:

G(e ^(jw))=N(ω)/D(ω)=jβ(ω))/(σ(ω)+jτ(ω))   (3)

If we set a_(k)=a_(k) ^(r)+ja_(k) ^(i) and b_(k)=b_(k) ^(r)+jb_(k) ^(j)we represent the desired Transfer Function of the IIR filter 146 to beF(e^(jω)) then,

F(e ^(jω))=R(e ^(jω))+jI(e ^(jω))   (4)

If the Error between the desired transfer function and the actualtransfer function is represented by ξ(ω)

Then, μ(ω)=F(ω)−G(ω)=F(ω)−N(ω))/D(ω))   (5)

and ξ(ω)D(ω)=F(ω)D(ω)−N(ω)   (6)

Further,

α(ω)=a ₀ ^(r)+(a ₁ ^(r) cos ω+a ₁ ^(j) sin ω)+ . . . +(a _(p) ^(r) cospω+a _(p) ^(j) sin pω)   (6)

β(ω)=a ₀ ^(j)+(a ₁ ^(j) cos ω−a ₁ ^(r) sin ω)+ . . . +(a _(p) ^(r) cospω−a _(p) ^(r) sin pω)   (7)

σ(ω)=1+(b ₁ ^(r) cos ω+b ₁ ^(j) sin ω)+ . . . +(b _(q) ^(r) cos qω+b_(q) ^(j) sin qω)   (8)

τ(ω)=(b ₁ ^(j) cos ω−b ₁ ^(r) sin ω)+ . . . +(b _(q) ^(j) cos qω−b _(q)^(r) sin qω)   (9)

It further follows that Dξ=(R+jI)(σ+jτ)−(α+jβ)   (10)

or,

$\begin{matrix}{{D\; \xi} = {{R\; \sigma} - {I\; \tau} + {j\left( {I\; \sigma \mspace{14mu} R\; \tau} \right)} - \alpha - {{j\beta}{or}k}}} & (11) \\{{D\; \xi} = {{\left( {{R\; \sigma} - {I\; \tau} - \alpha} \right) + {j\left( {{I\; \sigma} + {R\; \tau} - \beta} \right)}} = {{A(\omega)} + {{j\beta}(\omega)}}}} & (13)\end{matrix}$

At discrete angular frequencies the magnitude squared of Dξ is then

|D(ω_(k))ξ(ω_(k))|² =A ²(ω_(k))+B ²(ω_(k))   (14)

The total error E is the error obtained from summing the errors at a setof m discrete frequencies.

$\begin{matrix}{E = {{\sum\limits_{k = 0}^{m}{{{D\left( \omega_{k} \right)}{\xi \left( \omega_{k} \right)}}}^{2}} = {\sum\limits_{k = 0}^{m}\left\lbrack {{A^{2}\left( \omega_{k} \right)} + {B^{2}\left( \omega_{k} \right)}} \right\rbrack}}} & (15)\end{matrix}$

Thus,

m E=Σ[(R _(k)σ_(k) −I _(k)τ_(k)−α_(k))²+(I _(k)σ_(k) +R_(k)τ_(k)−β_(k))²]  (16)

If the partial derivative of E with respect to a_(i) ^(r) is taken, theerror is minimized if this derivative is set to 0.

$\begin{matrix}{{{\partial E}/{\partial a_{i}^{r}}} = {\sum\limits_{k = 0}^{m}\left\lbrack {{{- 2}\left( {{r_{k}\sigma_{k}} - {I_{k}\tau_{k}} - \alpha_{k}} \right){{\partial\alpha_{k}}/{\partial a_{i}^{r}}}} - {2\left( {{I_{k}\sigma_{k}} + {R_{k}\tau_{k}} - \beta_{k}} \right){{\partial\beta_{k}}/{\partial a_{i}^{r}}}}} \right\rbrack}} & (17)\end{matrix}$

resulting in Equation (18) below:

$\begin{matrix}\left. {{{\partial E}/{\partial a_{i}^{r}}} = {{\sum\limits_{k = 0}^{m}{\left\lbrack {{R_{k}\sigma_{k}} - {I_{k}\tau_{k}} - \alpha_{k}} \right){\cos \left( {\; \omega_{k}} \right)}}} - {\left( {{I_{k}\sigma_{k}} + {R_{k}\tau_{k}} - \beta_{k}} \right){\sin \left( {\; \omega_{k}} \right)}}}} \right\rbrack & (18)\end{matrix}$

Similarly, if the partial derivative of the Error with respect to α₁^(j), b_(i) ^(r), and b₁ ^(j) are taken the following equations (19),(20), and (21) respectively result.

$\begin{matrix}{{{\partial E}/a_{i}^{j}} = {0 = {\sum\limits^{m}\left\lbrack {{{- \left( {{R_{k}\sigma_{k}} - {I_{k}\tau_{k}} - \alpha_{k}} \right)}{\sin \left( {\; \omega_{k}} \right)}} - {\left( {{I_{k}\sigma_{k}} + {R_{k}\tau_{k}} - \beta_{k}} \right){\cos \left( {\; \omega_{k}} \right)}}} \right\rbrack}}} & (19) \\\left. {{{\partial E}/{\partial b_{i}^{r}}} = {0 = {{\overset{m}{\sum\limits_{k = 0}}\left\lbrack {{\left( {{R_{k}\sigma_{k}} - {I_{k}\tau_{k}} - \alpha_{k}} \right)\left( {R_{k}{\cos \left( {\; \omega_{k}} \right)}} \right)} + {I_{k}{\sin \left( {\; \omega_{k}} \right)}}} \right)} + {\left( {{I_{k}\sigma_{k}} + {R_{k}\tau_{k}} - \beta_{k}} \right)\left( {{I_{k}{\cos \left( {\; \omega_{k}} \right)}} - {R_{k}{\sin \left( {\; \omega_{k}} \right)}}} \right)}}}} \right\rbrack & (20) \\{\mspace{79mu} {and}} & \; \\\left. {{{\partial E}/{\partial b_{i}^{j}}} = {0 = {{\overset{m}{\sum\limits_{k = 0}}\left\lbrack {{\left( {{R_{k}\sigma_{k}} - {I_{k}\tau_{k}} - \alpha_{k}} \right)\left( {R_{k}{\sin \left( {\; \omega_{k}} \right)}} \right)} - {I_{k}{\cos \left( {\; \omega_{k}} \right)}}} \right)} + {\left( {{I_{k}\sigma_{k}} + {R_{k}\tau_{k}} - \beta_{k}} \right)\left( {{I_{k}{\sin \left( {\; \omega_{k}} \right)}} + {R_{k}{\cos \left( {\; \omega_{k}} \right)}}} \right)}}}} \right\rbrack & (21)\end{matrix}$

Using known matrix techniques for solving simultaneous equations thevalues of a_(i) and b_(j) for the IIR filter 146 are obtained whichminimize the error between an output of the IIR filter 146 and themeasured self-interference channel characterization. Advantageously,these calculations are done periodically, dependent on the operationalstate of the system 130, in order to update the IIR filter 146parameters as the self-interference channel changes.

A least mean squares curve fitting algorithm is applied by the leastsquares estimator 147 in order to find a_(m) and b_(n) such that E, thesum of the absolute values of the squares of the difference between theresponse of the IIR filter 146 and the measured self-interferencechannel estimate, is minimized.

E=Σ∥H(e ^(jω) _(k))−H _(k)∥², for all values of k (0−k)   (22)

Best fit of the IIR filter 146 values for a_(i) and b_(i) when equations(19), (20), (21) and (22) above are minimized by the processor in thecontroller 121. In an exemplary operational case, where p=q=12, fiftysimultaneous equations result, 26 for zeroes and 24 for poles. Thesemultiple equations can be expressed in terms of a matrix which can besolved using known techniques. This same procedure can then be followedfor other possible combinations of numbers of poles and zeros in orderto find the number of poles and zeros that minimize the error.

As real world channels between transmitter and receiver in the system 98and the system 130 within a full-duplex transceiver are subject toconstant change during the operation of the system 130 and the system 98and in certain self-interference channel scenarios the actual channelresponse can exhibit scenarios where there are large losses at one ormore points across the receive band and/or one or more large gainsacross the receive band, the actual number of poles and/or zeroes of theIIR filter 146 can be changed during the operation to provide the bestfit between the actual self-interference channel and the characterizedself-interference channel. As discussed, dependent on the state of theself-interference channel during an instance of the operation of thesystem 130, some value(s) or one or more ai and/or bi may be zero whichmay be optimal and represent the best fit for the given parameters ofthe IIR filter 146 for the given state of the self-interference channel.The full-duplex transceiver can be located in a base station (e.g., acommunications substation), a handset (e.g., a mobile handset), asatellite, etc.

Least-Squares Algorithm. As described in the equations above for solvingcoefficients for a given filter structure of the IIR filter 146. Theequations around how the best number of poles and zeros isalgorithmically determined by the following steps:

1. Start with the maximum number of poles and zeros.

2. Compute the filter coefficients of the filter 146 that give theleast-squares approximation to the measured filter coefficients (usingthe equations above).

3. Compute a residual error for this number of poles and zeros. (E fromabove).

4. Check that the poles are all within a radius of less than 0.99. Thisensures that the filter is stable and can be realized easily (i.e., notmarginally stable or unstable).

5. Repeat steps 2-4 with one less number of zeros each time until thenumber of zeros is 0 (i.e., only poles). Out of all of these filters,pick the one with the lowest residual error.

6. Set the number of zeros to maximum again. Repeat steps 2-5 with oneless number of poles each time until the number of poles is 0 (i.e., anFIR filter). Out of all of these filters, pick the one with the lowestresidual error.

Further to the stability of the IIR filter 146, the IIR filter 146 isstable if all of its poles lie within the unit circle (i.e., theirmagnitude is less than 1). This can be understood by looking at thetime-domain representation of a pole zp. In the time domain, a pole atzp of the IIR filter 146 has the form h[n]=sum(zp^(n)). If |zp|<1, then|zp^(n)|->0 for large n, and the series converges. If |zp|>=1, then theseries diverges. The placement of zeros does not affect stability (i.e.,when the IIR filter 146 has zeros outside the unit circle, the IIRfilter 146 is stable), but it does affect the response of the IIR filter146.

For a given number of poles (q) and zeros (p) of the IIR filter 146,there are 2^((p+1+q)) equations. This is because the numerator has onemore coefficient than the denominator and all of the coefficients arecomplex.

Implementations of the disclosure yielded good results for values ofp=q=12. It is recognized by those skilled in the art the higher thevalues of p and q, the more dynamic features can be matched correctly bythe IIR filter 146. Thus, the system 130 may be algorithmically adaptedor tuned for a given application of the disclosure.

The output of IIR filter 146 may be fed directly to position 135 alongwith Rx signal 131, which results in removing interference for Rx signal131, to create output 134 to receiver. The output of filter 146 is anequalized version of the interference present on Rx signal 131 input.The direct use of the IIR filter 146 output with no tone comparison andassociated components is preferably done when there is minimal concernwith regard to drift or gain control updates of Rx signal 131.

With continued reference to FIG. 5A, it should be understood that attimes automatic gain control and temperature may cause variationsbetween the two paths (between Rx signal 131 path and Rx signal 149path) that occur on a faster timescale than the measurement of thechannel and calculation of new IIR coefficients. Therefore, Fast Fouriertransform (FFT) block 133 and FFT block 140 may be used to measure themagnitude and phase of a fixed tone, such as the second fixed tone, oneach path. The magnitude and phase are then compared at comparison block136, and the results are fed into integrators in order to makeadjustments so that there is no DC offset between the two paths in bothmagnitude and phase. The outputs of these integrators (e.g., phase loopfilter 141 and gain loop filter 144) are then used to adjust the IIRfilter 146 output so that the output of phase shifter 143 has the samemagnitude and phase as the receive signal input (Rx signal 131). Theresulting signals are then subtracted at 135, which removes interferencefrom Rx signal 131 (the received signal). As shown in FIG. 5A, theoutput of the Variable Phase Shift element 143 is input to a complexmultiplier 166 which multiplies the output of the variable phase shiftelement 143 by a complex multiplication factor, as is further describedbelow, to shift the output of the Variable Phase Shift element 143 withrespect to both amplitude and phase. This is done before a comparison ismade at comparator 135 in order to scale the signal output by theVariable Phase Shift element 143 to optimize the effect of the scaledcompensation signal output by the complex multiplier 166. Here, thescaling is done with respect to both amplitude and phase using a complexscaling factor. The resulting signals are then subtracted at the adder135, which removes interference from Rx signal 131 (the receivedsignal).

FIG. 5D further illustrates the operation of the complex multiplier 166.Here, a representation of a magnitude on an axis 802 of H(e^(jω)) isshown for illustrative purposes. This shows another example of measuredamplitude circuit response as previously illustrated in FIGS. 8A and 8B.The magnitude scale on the axis 802 is in dB and a frequency scale on anaxis 804 is in normalized angular frequency 2πf. A similarrepresentation as previously also shown in FIGS. 8A and 8B would applyfor the phase of H(e^(jω)) but is not shown as the processes describedbelow for both magnitude and phase are similar.

A best fit curve is generated using Least Square functions in theleast-squares estimator 147 of FIG. 5A. The best fit curve is as a curve808. Two other representations of the best fit curve are shown as curves810 and 812. The curves 810 and 812 have exactly the same shape as thebest fit curve 808 with the exception that for the curve 812, theentirety of the curve 812 is translated to pass directly through ameasurement point 816, which has been made at the normalized frequencyof zero, i.e., at a middle of the band of interest. The curve 810 hasbeen translated down to pass directly through a point 814, which occursat the frequency of the fixed tone used for automatic gain control.Here, as further illustrated in FIG. 5D, if the curve 808 anchored atthe measurement point 816 measured at the center of the band of interestfor a given application of the disclosure, the error magnitude isillustratively shown by an error magnitude curve 820.

Illustratively, outside the region in the center of a curve 820indicating an error magnitude, the error magnitude is on the order of−15 db, while at the center of the curve 820, the error magnitude is onthe order of −35 db. Thus, for example, if the system 98 and the system130 were deployed in a narrowband air interface, error performance isillustratively better in the center of the band as is favored by thespectral shape of the signal. The bandwidth of the improved region aswell as the actual shape of the error magnitude curve is dependent onhow well the infinite impulse response of the IIR filter 146 of FIG. 5Amatches the actual circuit response of the self-interference channelformed between the transmitter 100 and the receiver 500 of a full-duplextransceiver.

Alternatively, if the curve 808 is anchored on a fixed tone 814, theerror magnitude 818 is as is illustratively also shown in FIG. 5D. Here,the error magnitude 818 shows improved error performance around thefixed tone 814 frequency. Again, outside the improved error performanceregion, the error magnitude is on the order of −15 db while the errorperformance can be improved to be on the order of −35 db. Further, thebandwidth of the improved region as well as the actual shape of theerror magnitude curve is dependent on how well the IIR filter 146generated matches the actual circuit response of the self-interferencechannel. Thus, as can be seen, the actual anchor point which may beoptimal for a given application may not be either point, and may beoptimal if the curve 808 is anchored at another point. A complexcorrection multiplier is generated according to the relationship)

H₀/H(e^(jω0))×H(e^(jωf))/H_(f)   (23)

Here, H₀ is actual measured value at the center of the band of interestat a normalized angular frequency of ω₀; H_(f) is the actual measuredvalue at a given angular frequency ω_(f); H(e^(jω0)) is the IIR filter146 response at the center of the band of interest and H(e^(jωf)) is theBR filter 146 response at a given frequency.

This correction multiplier is a complex multiplier (similar to thecomplex multiplier 125) accounts for both amplitude and phase. Not shownin FIG. 5A, the correction multiplier is computed in the controller 121or in the alternative, in another computing element (not shown in FIG.5A).

The scaling factor was chosen in this manner to offset the effects ofthe feedback loop formed by the FFT 140, the comparison block 136, thephase loop filter 141, and the gain loop filter 144 that uses the fixedtone 814. The feedback loop ensures that after the multiplier 145 andthe phase shifter 143 (also referred to as the phase shift element 143),the magnitude of the fixed tone 814 is the same on both paths at thefrequency of the tone. This complex scaling factor introduced by thecomplex multiplier 166 shifts that result from being true around thefixed tone 814 to instead be focused on the center of the band ofinterest.

Computation of the scaling factor is periodically done during theoperation of the system 130 and the system 98 as the circuit response isre-determined using self-interference channel characterization swepttone sweeps, re-computation of the channel characterization andre-computation of the least square circuit response at the least squaresestimator 147.

FIG. 5B illustrates an exemplary method that accounts for variations inthe received signal (Rx signal 131) and the output of the IIR filter146, which is based on the sample of the transmit signal (Rx signal149). At block 161, a magnitude and a phase of a tone in a receivesignal (e.g., Rx signal 131) is determined. The magnitude and phase maybe determined based on the use of a Fast Fourier Transform (FFT) or thelike. As illustrated in FIG. 5A, FFT 133 may be used to determine themagnitude and phase of a fixed tone in Rx signal 131. At block 162, amagnitude and a phase of a tone of a sampled transmit signal (e.g., Rxsignal 149) is determined. The sampled transmit signal may have beenthrough an IIR filter, such as IIR filter 146. For example, FFT 140 ofFIG. 5A may be used to determine the magnitude and phase of the fixedtone of the sampled transmit signal. At block 163, the magnitude andphase of the tone of the receive signal and the tone of the sampledtransmit signal may be compared. For example, comparison block 136 ofFIG. 5A may compare the magnitude and phase of the tones.

With continued reference to FIG. 5B, at block 164, the magnitude of thesampled transmit signal is adjusted based on the comparison of block163. For example, with reference to FIG. 5A, gain loop filter 144 may befed results of the comparison block 136 along segment 137. The output ofgain loop filter 144 is used to adjust IIR filter 146 output (which isbased on the sampled transmit signal) so that the output of the phaseshifter 143 has the appropriate magnitude to cancel interference in Rxsignal 131 (the receive signal). At block 165, the phase of the sampledtransmit signal is adjusted based on the comparison of block 163. Forexample, with reference to FIG. 5A, phase loop filter 141 may be fedresults of the comparison block 136 along segment 138. The output ofphase loop filter 141 is used to adjust IIR filter 146 output (which isbased on the sampled transmit signal) so that the output of the phaseshifter 143 has the appropriate phase to cancel interference in Rxsignal 131 (the receive signal).

In some situations digital filters, such as FIR or IIR operating atbaseband on the digital samples, may be used to remove noise before the“clean” sample of output 134. In other situations RF filters may beremoved as may be useful in some software-defined operational modes.

The final stage of the digital self-interference residual cancellationis a set of feedback loops that keep the system stable between theequalizations mentioned above. The loops, such as the loop with phaseloop filter 141 and the loop with gain loop filter 144, regulate themagnitude and phase of the sampled signal so that when the Rx signal 131and Rx signal 149 are subtracted, the power in the fixed tone (e.g., thesecond fixed tone) used for automatic gain control (AGC) is minimized.

For additional perspective, a discussion of automatic gain control (AGC)as associated with digital self-interference residual cancellation isprovided below. Since the two receivers have independent AGC loops,there are conceivable situations where the AGC on one receiver willchange suddenly. The feedback loops, such as the loop with phase loopfilter 141 and the loop with gain loop filter 144, in the system thattry to match the magnitude and phase of a second fixed tone between thetwo paths are designed to be fast enough that they can compensate forthe sudden change in gain that would result from an AGC update. Notethat in this case, each receiver is assumed to have independent AGCcontrol, so no coordination is assumed between them. This allows eachreceiver to compensate for sudden power changes at its input. It isexpected that Rx signal 149 AGC will remain relatively stable since itis measuring a known signal sampled from a power amplifier output (e.g.,power amplifier 104 of FIG. 1B).

As discussed herein, the compensated signal based on the output of block132 and block 143 of FIG. 5A of the two feedback loops is subtractedfrom the receive signal (Rx signal 131). The result of this subtractionis the output of the noise canceller and contains output 134 to receiverin which transmit residual noise interference is greatly reduced inpower. Here, with regard to the output 134, the transmit interference inthe receive band has been cancelled or attenuated by some amount.

Controller 121 may exert control over the elements of system 130.Illustratively, in implementations of system 130, coefficients and otherdata that apply in a given mode of operation of system 130 can becalibrated and stored for invocation as needed throughout the operationof system 130 for given modes of operation during given instants ofoperation of system 130. Thus, multi-band, multi-radio access technologyoperational modes can be changed by controller 121 as needed. Thisenables, for example, high security operation where the mode ofoperation of the system including bands can be changed on the fly.

Self-Interference Channel Estimation Example

FIG. 5C illustrates a flowchart illustrating operations of a method 2100for characterizing a self-interference channel of a full-duplextransceiver including the transmitter 100 and a receiver 500. The method2100 may begin in an operation 2102 in which the fixed tone and theswept tone are generated at the transmitter 100 of the full-duplextransceiver. The fixed tone and the swept tone may be digitallygenerated by the controller 121 and then converted into correspondinganalog signals by a digital to analog converter at the transmitter 100.The fixed tone may be generated to have an analog frequency above anupper edge of the reception band of the receiver 500. The swept tone maybe generated to have a frequency ranging from a lower edge of thereception band to an upper edge of the reception band. The swept tonemay be generated periodically at discretely spaced frequencies withinthe reception band. The fixed tone and the swept tone may be injected toa message signal to be transmitted by the transmitter 100.

In an operation 2104, the controller 121 of the transmitter 100up-converts the fixed tone, the swept tone and the message signal toradio-frequency corresponding to the transmission band of thetransmitter 100. The up-converted signal is the transmission signal.

In an operation 2106, a sample of the transmission signal is provided tothe receiver 500, collocated with the transmitter 100. The sampleprovided to the receiver 500 is free of external noise and degradationdue to the self-interference channel formed between the transmitter 100and the receiver 500.

In an operation 2108, the transmission signal including the messagesignal, the fixed tone, and the swept tone that changes frequency frombelow an upper edge to above a lower edge of the reception band of thereceiver 500 is transmitted. The fixed tone is used for the fastfeedback loop used for AGC and temperature compensation.

In an operation 2110, at the IIR filter 146 of the receiver 500, aninfinite impulse response of the self-interference channel based upon areception of the swept tone swept at each frequency in the receptionband is determined using the equations 1-22 above.

In an operation 2112, the self-interference channel between thetransmitter 100 and the receiver 500 is estimated based upon thecoefficients of the IIR filter 146, as also discussed in the equations1-22 above. The self-interference is known from the transmit sample thatis fed into the system as the signal 149. The purpose of the BR filter146 is to equalize the frequency response that has been applied tosignal 131 and signal 149 so that they can be subtracted with anequivalent frequency response, leading to broadband cancellation.

Tone Based IQ Compensation Example

Referring to FIG. 5E, a method 2200 for in-phase and quadrature phase(IQ) compensation in a full duplex transceiver including the transmitter100 and the receiver 500 are illustrated. The method 2200 may begin inan operation 2202 in which a fixed tone at a frequency lower than alower edge of the reception band may be generated. The generation of thelower edge fixed tone may be carried out digitally by the controller 121in the transmitter 100 prior to any transmission of a message signal bythe transmitter 100.

In an operation 2204, the fixed tone and the message signal may beup-converted for transmission as a transmission signal in the radiofrequency domain. The up-conversion may be to a frequency oftransmission in the transmission band of the transmitter 100.

In an operation 2206, the transmission signal including the fixed toneand the message signal may be transmitted by the transmitter 100.

In an operation 2208, the receiver 500 may receive a portion of thetransmission signal including the fixed tone and the message signal as areception signal. In one aspect, the reception signal may bedown-converted to a lower or intermediate frequency before a processorin the controller 121 may process the reception signal. It will beappreciated that such down-conversion is part of a normal operation ofthe receiver as other received signals may be down-converted by thereceiver 500, in addition to the leaked transmission signal received asthe reception signal in the operation 2208.

In an operation 2210, the processor of the receiver 500 may determine again mismatch (g) and a phase mismatch (φ)) between an in-phase (I)component and a quadrature (Q) phase component of the reception signalby detecting an image tone of the fixed tone in the reception signal.

In an operation 2212, the image tone is minimized to compensate the gainmismatch and the phase mismatch between the I and Q components of thereceived signal by the processor. The minimization may includecomputing, at the processor, the FFT 140 of the reception signal (alsoreferred to as the received signal), outputting, at the processor, toindex into the FFT result to obtain the result in a particular bin(corresponding to the frequency of the fixed tone and its image), theFFT 140 by a first constant corresponding to the fixed tone to result ina first product and by a second constant corresponding to the image tonecorresponding to a second product, combining, at the processor, thefirst product and the second product to form an error signal, splitting,at the processor, the error signal into a real error component and animaginary error component, integrating, at the processor, the real errorcomponent and the imaginary error component to force a DC error in theerror signal to zero, splitting the reception signal into a realcomponent and an imaginary component, multiplying the real component bya first coefficient and the imaginary component into a secondcoefficient, adding the imaginary component after the multiplying by thesecond coefficient to the reception signal, and combining the imaginarycomponent in the reception signal after the adding to the real componentafter the multiplying by the first coefficient as an input to the FFT140.

In one aspect, the minimizing may be performed using a feedback loop inthe receiver, the feedback loop including the gain loop filter 144 forminimizing the gain mismatch and the phase loop filter 141 forminimizing the phase mismatch. In another aspect, the minimizing mayinclude determining, at the processor, a matrix E as E=[1 0; −g sin φ, gcos φ], determining, at the processor, an inverse E-1 of the matrix E,and applying the inverse E-1 to the received signal for thecompensating.

Simulations of system 130 have shown the degree of isolation improvementor interference cancellation that can be obtained with system 130. Theseresults have been verified against actual implementation of the system.The results of these simulations are based on the accuracy of phaseand/or amplitude error between the residual error compensation signaland the actual received signal. FIGS. 9A-D illustrate the degree ofimprovement in isolation which are obtainable for different combinationsof amplitude error or phase error. Thus, as is seen, implementations ofthe system may be optimized for given applications according toisolation needs.

FIG. 3 illustrates a tone generation function used as part of thedigital residual interference cancellation loop according to an aspectof this disclosure. Prior to transmission, a lower band edge fixed tone9, an in band swept tone 10 swept at discrete frequencies, and an upperband edge tone 11 may be combined and up-converted at an up-converter302.

Exemplary results of operation of the canceller are shown in FIGS. 4, 10and 11. FIG. 4 illustrates Rx signal 149 and Rx signal 131 for anexemplary 5 MHz LTE case. Signal 175 is the received signal (e.g., Rxsignal 131) containing a 5 MHz LTE signal, a Tx interference signal thatis roughly 20 dB above the thermal noise floor, and the tones used forcancellation. Signal 173 is a low-pass filtered version of thecompensated sampled signal (e.g., Rx signal 149 path after the variablephase shift 143 in FIG. 5A). During normal operation, the tones in theRx channel will not be present unless an equalization operation isoccurring. Signal 174 of FIG. 4 shows the output a digitalself-interference residual cancellation system (e.g., output 134 ofsystem 130), which illustrates the previously buried LTE signal and theexpected 20 dB of interference cancellation to the noise floor of thereceiver. The ability to cancel all the way to the thermal floor of thereceive means that this digital self-interference residual cancellationsystem can be used with no degradation of the radio's noise figure. Theradio's noise figure is the degradation of the signal to noise ratio ofthe system. If there is an inability to cancel to the thermal noisefloor, the sensitivity would be less than optimal, so that could bedescribed as either lower sensitivity within the radio or a highersystem noise figure for the radio.

FIG. 10 shows an operation where some interference cancellation hasoccurred using feed-forward self-interference cancellation, but withoutapplication of the digital self-interference residual cancellation asdisclosed herein. Multiple representations of the de-modulated 5 MHz LTEwaveform before digital interference cancellation are shown in FIG. 10.In block 181, the signal constellation is shown, while in block 182,OFDM Error Vector Spectrum is shown. At block 183, the spectrum is shownand at block 184 a summary of error elements is shown. Since residualnoise has not been cancelled, the constellation is collapsed and thedata are unrecoverable.

FIG. 11 shows operation with both the interference cancellation usingfeed-forward self-interference cancellation and digitalself-interference residual cancellation is applied. Here, multiplerepresentations of the de-modulated waveform after digital interferencecancellation are shown as in FIG. 10 when digital interferencecancellation is not applied. Block 186 of FIG. 11 shows the signalconstellation, block 187 shows the OFDM Error Vector Spectrum, block 188shows the spectrum, and block 189 shows a summary of error elements. Thesignal is recovered from far beneath the interference. Error vectormagnitude is marginally affected due to the presence of the tones,however, this is a temporary condition as channel characterization tonesare only swept across the band periodically. As shown, however, even inthe presence of channel characterization tones the Error Vectorperformance is good.

Self-Interference Cancellation Antenna Systems and Methods

FIG. 12A illustrates a self-interference cancellation system accordingto an aspect of the disclosure. In FIG. 12A, a feed-forwardself-interference cancellation loop (FIG. 1A) and a digital residualinterference cancellation loop (FIG. 2) are connected in tandem. In anembodiment, the transmit signal 1202 sourced by modulation and basebandprocessing functions may be input to the feed-forward self-interferencecancellation loop 1204. The feed-forward self-interference cancellationloop 1204 may include a power amplifier. In certain embodiments, thehigh power signal 1210 may include residual noise after carriercancellation the feed-forward self-interference cancellation loop 1204and input to the digital residual interference cancellation loop 1206.The high power transmit antenna signal 1212 may either be fed to atransmit antenna or to further isolation mechanisms (as seen in FIGS.13-16). In some embodiments, the receive signal 1214 may be eithersourced by a receive antenna or further isolation mechanisms (e.g.,FIGS. 13-16). In the present disclosure, the isolation obtainable usingthe two cancellation loops, the feed-forward self-interferencecancellation loop 1204 and input to the digital residual interferencecancellation loop 1206, may be further improved by application ofantenna isolation methods.

FIG. 12B illustrates a self-interference cancellation system accordingto another aspect of the disclosure. The embodiment in FIG. 12B includesthe subsystems of the feed-forward self-interference cancellation loop1204 (from FIG. 1), the digital residual interference cancellation loop1206 (from FIG. 2), and a self-interference cancellation circuit 1216.In other embodiments, the system 1250 may include an additional circuit1218. The additional circuit 1218 may include other antenna isolationmethods, such as using a separate antenna 1260 to receive a signal, ortransmitting the transmitted signal at a different polarization from thereceived signal, for example.

The system 1250 may operate on any wireless communication standard suchas Global Systems for Mobile communications (GSM), Universal MobileTelecommunications System (UMTS), or Long Term Evolution (LTE), and maybe configured to handle as many frequency bands as needed. In anembodiment, for example, the system 1250 may handle 4 differentfrequency bands. The system 1250 may be sized as needed, such as, in anexample embodiment where the system 1250 can handle 4 bands, thefeed-forward self-interference cancellation loop 1204 may be about 2.25in², the digital residual interference cancellation loop 1206 may beabout 5.75 in², and the self-interference cancellation circuit 1216 maybe about 4.75 in². Of course, in other embodiments, other dimensions maybe used.

The system 1250 may be configured to provide any amount of attenuationof the self-interference necessary, typically on the order of about30-110 dB. In certain embodiments, different configurations of thesystem 1250 may be used or the components in the individual subsystemsmay be varied to arrive at different amounts of attenuation. Forexample, in an embodiment the feed-forward self-interferencecancellation loop 1204 and the digital residual interferencecancellation loop 1206 may be used to provide a desired amount ofattenuation without additional attenuation circuitry. In such anexample, the feed-forward self-interference cancellation loop 1204 mayprovide about 40 dB of attenuation, while the digital residualinterference cancellation loop 1206 may provide about 30 dB ofattenuation, resulting in about a 70 dB of attenuation in the system1250. In other embodiments, other combinations of the subsystems, orcomponents or methods in each subsystem may be used to provide thedesired attenuation. The subsystems may also each have an insertion lossassociated with them, for example, the feed-forward self-interferencecancellation loop 1204 and the digital residual interferencecancellation loop 1206 may each have an insertion loss of about 0.2 dB,and the self-interference cancellation circuit 1216 may have aninsertion loss of about 0.5 dB.

FIG. 13A illustrates diagram of a self-interference cancellation circuitwith two circulators, a phase shifter, and a filter according to anaspect of the disclosure. The use of such a self-interferencecancellation circuit may remove a high power carrier from a receivechain while cancelling noise in the receive band due to the transmitter.The system may use circulators in filtering applications in several waysto allow cheaper and less complex filters to be used, and aid in thecancellation of reflected transmit noise from an antenna.

In the circuit 1300 of FIG. 13A, a high power carrier may be removedfrom the receive path while cancelling noise in the receive band due tothe transmitter. In an embodiment, a transmitter power amplifier 1304may be configured to amplify a transmit signal 1302. The amplifiedtransmit signal 1302 may then be sent to a first circulator 1306, wherethe first circulator 1306 may be coupled to the transmitter poweramplifier 1304 through a port 1. In an embodiment, circulator 1306 mayhave an insertion loss of about 0.1-0.2 dB between port 1 and port 2. Asecond circulator 1308 may be coupled through a port 1 of the secondcirculator 1308 to the port 2 of first circulator 1306. In someembodiments, an antenna 1310 may be coupled to a port 2 of the secondcirculator 1308. The antenna 1310 may be located at an end of thetransmission-reception path, where the antenna 1310 may transmit thetransmit signal 1302 and receive a receive signal 1324. The transmitpower of the antenna 1310 may be any power suitable to the application,for example, the transmit power may be about 40 dBm. In an exampleembodiment, the return loss of the antenna 1310 may be about 10-15 dB.

In certain embodiments, the first circulator 1306 and second circulator1308 may be aligned along a transmission-reception path of theself-interference cancellation circuit 1300. A receiver amplifier 1322may be coupled to a reception path of the self-interference cancellationsystem, where receive signal 1324 may be output to digital residualinterference cancellation loop 1206 (FIG. 12A) or a receiver. It iscontemplated that in some embodiments, the receiver amplifier 1322 maybe coupled to a port 3 of first circulator 1306. The typical power fromport 1 to port 3 of circulator 1306 may be about 15-25 dB less than thatof the incident transmit power. In an embodiment, there may be about 20dB of isolation between port 1 and port 3 of circulator 1306.

A phase shifter 1312 may be located on the reception path and coupled toa port 3 of the second circulator 1308. In certain embodiments, thephase shifter 1312 may be adjusted to shift the phase of a signalpassing through it a desired number of degrees such that the signal canbe combined with the signal after port 3 of circulator 1306 to cancelthe transmit noise in the receive band. In a circuit such as in FIG.13A, where the signal passes through the phase shifter 1312 twice, onceon its path from the transmitter, and once as it is reflected off thefilter 1316, the phase shifter 1312 may shift the phase of the signal byhalf of the total amount desired for cancellation each time the signalpasses through the phase shifter 1312. For example, if 30 degrees ofphase shift is needed for cancellation, the phase shifter 1312 may shiftthe signal 15 degrees each time the signal passes through, resulting ina total of 30 degrees of phase shift.

In an embodiment shown in FIG. 13A, a variable attenuator 1314 may becoupled to filter 1316 and the phase shifter 1312. The variableattenuator 1314 may adjust the amplitude of the signal passing throughit as needed, for example, to match the power of reflected noise fromthe antenna 1310 with the power of the transmit noise in the receiveband from the transmitter in order to cancel the noise before it reachesthe receiver. In a circuit such as in FIG. 13A, where the signal passesthrough the variable attenuator 1314 twice, once on it path from thetransmitter, and once as it is reflected off the filter 1316, thevariable attenuator 1314 may attenuate the signal by half of the totalamount desired each time the signal passes through the variableattenuator 1314. For example, if 10 dB of attenuation is needed, thevariable attenuator 1314 may attenuate the signal 5 dB each time thesignal passes through, resulting in a total of 10 dB of attenuation. Inother embodiments, the variable attenuator 1314 may be a fixedattenuator. The filter 1316 may be located between the variableattenuator 1314 and a load 1318, where load 1318 may be coupled toelectrical ground 1320. In embodiments, the filter 1316 may be anyfilter suitable to the application, such as a transmitter bandpassfilter or a receiver notch filter, for example. In other embodiments,the attenuator 1314, phase shifter 1312, and filter 1316 can be in anyorder.

The circuit 1300 of FIG. 13A may provide for the cancellation of thetransmit carrier in the receive path and the cancellation of transmitnoise in the receive band. A transmit carrier, along withintermodulation products may be injected at port 1 of circulator 1306.The signal may travel through both circulators 1306 and 1308 to theantenna 1310, where although most of the energy may be radiated, some ofthe energy may be reflected back towards circulator 1308. In anembodiment, the carrier may be directed into a load 1318, so that verylittle energy of the carrier can be reflected back to the circulator1308 and consequently, into the receive amplifier 1322. Energy in thereceive band can be reflected off of the filter 1316 and travel backthrough second circulator 1308, through first circulator 1306, and thento the receive amplifier 1322.

In some embodiments, the signal going into port 1 of circulator 1306 cantake alternate paths through the isolation of the circulators 1306 and1308, which can cause signal addition or cancellation. For example, whenthe transmit signal approaches circulator 1306, some of the signal mayleak from port 1 to port 3, allowing some of the transmit noise in thereceive band to enter the sensitive receiver. However, if the filteringis adjusted appropriately, then the noise reflected off of the antennacan be shifted so that when it reflects off of the filter 1316, andtravels from port 3 to port 1 of circulator 1308, and then from port 2to port 3 on circulator 1306, it can cancel with the transmit noise inthe receive band from the transmitter.

The system in FIG. 13A may provide several advantages over othersystems. For example, a high power filter may not be required in thetransmit path, thus reducing losses and increases overall efficiency.The filter 1316 after the second circulator 1308 can be a much lowerpower, because the reflected signal from the antenna 1310 may besignificantly less than the incident power to the antenna 1310. Inaddition, the transmit noise in the receive band can be canceled. Therejection of the noise in the receive path may be limited though, to theisolation of the circulators 1306 and 1308 against the return loss ofthe antenna 1310, and the tuning of the filter 1316. It is contemplatedthat the bandwidth of cancelation may also be dependent on theequivalent line length (e.g., phase delay) between each of the devicesalong the path from circulator 1306 to antenna 1310 and from circulator1306 to ground 1320. The rejection of the transmit carrier may belimited to the return loss of circulator 1306, since the carrier powerreflected from the antenna 1310 may be terminated with the filter 1316.In some embodiments, a low loss, low power filter (not shown) can beinserted in the receive path to remove the rest of the transmit carriersince the typical power from port 1 to port 3 of circulator 1306 may beabout 15-25 dB less than that of the incident transmit power. Thecircuit 1300 may be included in a wireless communication device such asa base station or a mobile phone.

FIG. 13B illustrates a method 1350 of self-interference cancellationaccording to an aspect of the disclosure. In step 1352, a transmitsignal may be generated along a transmit path of a transceiver. Thetransmit signal 1302 may then be sent through a circulator 1306 (seeFIG. 13A) to substantially isolate the transmit signal 1302 from areceiver, wherein at least a portion of the transmit signal 1302 entersa receive path towards the receiver in step 1354. At step 1356, thetransmit signal 1302 may be transmitted from an antenna 1310. In anembodiment, a signal 1324 may be reflected from the antenna 1310,wherein the reflected signal 1324 may be at substantially less powerthan an incident power to the antenna 1310 in step 1358. The reflectedsignal 1324 can include a transmitter carrier signal and a transmitternoise. In step 1360, a received signal 1324 may be routed from theantenna 1310. The reflected signal 1324 may be routed through a filter1316 in step 1362. In an exemplary embodiment in step 1364, thereflected and phase shifted transmitter noise may be combined with thereceived signal 1324 in the receive path to cancel the portion of thetransmit signal 1302 that entered the receive path towards the receiverfrom the circulator 1306.

FIG. 14 illustrates a diagram of a self-interference cancellationcircuit with two circulators, a phase shifter, and a filter according toanother aspect of the disclosure. In the circuit 1400 of FIG. 14, a highpower carrier may be removed from the receive path while cancellingnoise in the receive band due to the transmitter. The circuit 1400 mayhave a reduced line length in the signal path as compared to otherembodiments (e.g. FIG. 15), which may enhance the cancellation possible,and may provide a lower insertion loss in the transmit path.

In the circuit shown in FIG. 14, a transmit signal 1402 may be sent to atransmitter power amplifier 1404 configured to amplify the transmitsignal 1402, and send the transmit signal 1402 to a first circulator1406 through a port 1. A second circulator 1408 may be coupled through aport 1 of the second circulator 1408 to a port 3 of first circulator1406. In an embodiment, an antenna 1410 may be coupled to a port 2 ofthe first circulator 1406. The antenna 1410 may be located at an end ofthe transmission-reception path, where the antenna 1410 may transmit thetransmit signal 1402 and receive a receive signal 1424.

In certain embodiments, a receiver amplifier 1422 may be coupled to areception path of the self-interference cancellation circuit, wherereceive signal 1424 may be output to digital residual interferencecancellation loop 1206 (FIG. 12A) or a receiver. It is contemplated thatin some embodiments, the receiver amplifier 1422 may be coupled to aport 3 of second circulator 1406. This configuration may allow foradditional isolation of the transmit signal 1402 that may leak from port1 to port 3 of circulator 1406, since the leaked portions of thetransmit signal 1402 then have to pass through circulator 1408 before itcan enter the receiver. In an embodiment, the leaked portions of thetransmit signal 1402 may pass through filter 1416 to load 1418 thusfurther reducing noise that may be sent to the receiver.

A phase shifter 1412 may be located on the reception path and coupled toa port 2 of the second circulator 1408. In an embodiment shown in FIG.14, a variable attenuator 1414 may be coupled to filter 1416 and thephase shifter 1412. The filter 1416 may be located between the variableattenuator 1414 and a load 1418, where load 1418 is coupled toelectrical ground 1420. The filter 1416 may be any filter suitable tothe application, such as, for example, a transmitter bandpass filter ora receiver notch filter. In other embodiments, the attenuator 1414,phase shifter 1412, and filter 1416 can be in any order.

In certain embodiments, a controller 1426 may be coupled to theself-interference cancellation circuit 1400 to control the circuit. Thecontroller 1426 may be coupled to the circuit in any suitable manner tocontrol desired components of the circuit. For example, in FIG. 14, thecontroller 1426 may be coupled to the phase shifter 1412 and thevariable attenuator 1414, such that these components may be controlled.The controller 1426 may also be coupled to the circuit after the receiveamplifier 1422 to receive feedback information on the properties of thesignal to determine if adjustments to the other components such as thephase shifter 1412 and the variable attenuator 1414 may be needed. Afeedback and control algorithm may be applied by the controller 1426 toadjust the phase shifter 1412 and the attenuator 1414.

FIG. 15 illustrates diagram of a circuit with two circulators and afilter according to an aspect of the disclosure. In the circuit 1500 ofFIG. 15, a high power carrier may be removed from the receive path.

In an embodiment, a transmitter power amplifier 1504 may be configuredto amplify a transmit signal 1502. The amplified transmit signal 1502may then be sent to a first circulator 1506, where the first circulator1506 may be coupled to the transmitter power amplifier 1504 through aport 1 of the first circulator 1506. A second circulator 1508 may becoupled through a port 1 of the second circulator 1508 to a port 2 offirst circulator 1506. In an embodiment, an antenna 1510 may be coupledto a port 2 of the second circulator 1308. The antenna 1510 may belocated at an end of the transmission-reception path, where the antenna1510 may transmit the transmit signal 1502 and receive a receive signal1524.

In certain embodiments, the first circulator 1506 and second circulator1508 may be aligned along a transmission-reception path of the circuit1500. A receiver amplifier 1522 may be coupled to a reception path ofthe circuit, where receive signal 1524 may be output to the digitalresidual interference cancellation loop 1206 (FIG. 12A) or a receiver.It is contemplated that in some embodiments, the receiver amplifier 1522may be coupled to a port 3 of first circulator 1506.

A filter 1516 may be located on the reception path and coupled to a port3 of the second circulator 1508. In other embodiments, a variableattenuator may be coupled to filter 1516. The filter 1516 may be locatedbetween the second circulator 1508 and a load 1518, where load 1518 iscoupled to electrical ground 1520. In certain embodiments, the filter1516 may be any filter suitable to the application, such as atransmitter bandpass filter or a receiver notch filter, for example.

In an embodiment, the user can take advantage of the qualities of theantenna to reduce the power handling requirement of the filter. In FIG.15, a high power transmit signal 1502 may be present at the output ofthe transmit power amplifier 1504. This signal 1502 may pass throughfirst circulator 1506 and second circulator 1508 towards the antenna1510. The antenna 1510 may reflect some of the high power transmitsignal 1502 back due to the return loss of the antenna 1510, as well assend the receive signal 1524. Both signals 1502 and 1524 may then travelthrough the second circulator 1508 towards the filter 1516 and the load1518. In an embodiment where the filter 1516 is a transmit bandpassfilter, the filter 1526 can allow the reflected signal in the transmitband to go through to the load 1518 and be absorbed, while the receivedsignal may be reflected and return towards the second circulator 1508.The received signal can then proceed back to the first circulator 1506and then into the receiver amplifier 1522 where the signal may beamplified and sent to the receiver. In such an embodiment, the filter1526 may have a lower power handling requirement as a result of thereturn loss of the antenna 1510. In another embodiment, the filter 1526may be a receive notch filter.

An advantage of the circuit 1500 is that a notch filter or bandpass canbe easier to design for high power handling than a diplexer or otherfiltering methods. The system noise figure in the circuit 1500 may alsobe reasonably low, and can depend on the return loss of the antenna, thereturn loss of the circulators 1506 and 1508, the return loss of thefilter 1516 in the receive band, and the noise figure of the receiveramplifier 1522. In some embodiments, advantages in power handling of thefilter 1516 can be up to about 20 dB, but can be dependent on the returnloss of the antenna 1510. Circuit 1500 may also greatly attenuate thereflected transmit signal. The amount of attenuation can depend on thereturn loss of the filter 1516 used and the quality of the load 1518,and of course the quality of the matching of the components in the restof the circuit.

FIG. 16 illustrates diagram of a circuit with one circulators and aphase shifter according to an aspect of the disclosure. In the circuit1600 of FIG. 16, transmit noise present in the receive path may becanceled via phase shifter 1612. In an embodiment, a transmitter poweramplifier 1604 may be configured to amplify a transmit signal 1602. Theamplified transmit signal 1602 may then be sent to a first circulator1606, where the first circulator 1606 may be coupled to the transmitterpower amplifier 1604 through a port 1 of the first circulator 1606. Inthe embodiment shown in FIG. 16, the phase shifter 1612 may be locatedon the transmission-reception path and coupled to a port 2 of the firstcirculator 1606. Antenna 1610 may be located at an end of thetransmission-reception path and coupled to the phase shifter 1612. In anembodiment, the antenna 1610 may transmit the transmit signal 1602 andreceive a receive signal 1624.

A receiver amplifier 1622 may be coupled to a reception path of theself-interference cancellation system, where receive signal 1624 may beoutput to digital residual interference cancellation loop 1206 (FIG.12A) or a receiver. It is contemplated that in some embodiments, thereceiver amplifier 1622 may be coupled to a port 3 of first circulator1306.

In an embodiment where the power reflected from the antenna 1610 issimilar to the power leaked from port 1 to port 3 on the circulator1606, it may be possible to use a phase shifter 1612 in the path of theantenna 1610 to set the phase of the noise signal so that it cancels atport 3 of the circulator 1606. This may allow the cancellation of eitherthe transmit carrier power or the power amplifier noise in the receiveband depending on the phase setting.

Software Defined Radio Front End and Secure Radio Methods

According to another aspect of the disclosure, the adaptability of theart taught herein can be applied for the realization of high securitysystems. For example, one such application of the art of this disclosureis a more secure and robust frequency hopping method. Frequency hoppingis understood to increase security for communications. In frequencyhopping, the transmitter and receiver change the frequency at which theyare operating in a manner that is known to both the transmitter and thereceiver. FDD systems utilize multiple, non-programmable band passfilters. The band pass filters limited the usefulness of frequencyhopping because, due to, for example, space and cost constraints,transmitters and receivers could have a limited number of band passfilters. Thus, the number of frequency bands, and radio accesstechnologies, the transmitter and receiver could hop through waslimited.

As shown in FIG. 17, during the operation of the system, transmit andreceive operations of the system may be changed during the operation ofthe system to increase security to very high levels. For example, thismay be accomplished by removing the multiple, non-programmable band passfilters of the prior art. Instead, the system may use asoftware-implemented filter to process the transmit and receive signals.This disclosure may use a software-implemented filter rather thanmultiple, non-programmable band pass filters because of the feed-forwardand digital interference cancellation methods described above. Thesystem of this disclosure may use software-implemented filters becausethe feed-forward and digital cancellation methods remove transmit energythat would otherwise be present in the receive frequency band. Forexample, the band pass filters 3 and 13 may be implemented in software.As a result, the center frequency of the band being passed by the bandpass filters 3 and 13 may be changed to virtually any frequency suitablefor communications. Therefore, the feed-forward loop may be operable onany suitable Radio Access Technology as well. Additionally, oralternatively, the linear vector modulators 11 and 16 may also besoftware-controlled. For example, the amount the magnitude and phasevariation of the signals processed by the linear vector modulators 11and 16 may be software-controlled.

In the digital cancellation method, the fixed tones and swept tones maybe upconverted before being injected. The fixed and swept tones may beupconverted to the operating transmission and reception frequency of thetransceiver. If the operating transmission and reception frequency ofthe transceiver changes, as it would during frequency hopping, thefrequencies to which the fixed and swept tones may be converted maycorrespondingly change. Moreover, the magnitude of the fixed and swepttones may be software-controllable. Additionally, the coefficients forthe IIR filter 16 may be software-controlled as well.

As shown in FIG. 17, the transmitter may operate in, for example, threetransmit frequency bands 1902, 1904, 1906. During transmission, thetransmitter may transmit data on frequency band 1902. To increasesecurity of the transmissions, the transmitter may begin to transmitdata on a different frequency from frequency band 1902. For example, thetransmitter may switch from transmitting data on frequency band 1902 totransmitting data on frequency band 1906, as indicated by 1914. Thetransmitter may change which frequency band it is transmitting on basedon, for example, a programmed frequency hopping algorithm. Aftertransmitting data on frequency band 1906, the transmitter may againchange the frequency band on which it operates. For example, thetransmitter may switch from operating on frequency band 1906 tooperating on frequency band 1904, as indicated by 1916. Thereafter, thetransmitter may again switch the frequency band on which it istransmitting. For example, as indicated by 1918, the transmitter mayswitch from operating on frequency band 1904 and return to operating onfrequency band 1906. The transmitter, after transmitting on frequencyband 1906 again, may switch the frequency band on which it operates. Forexample, the transmitter may start transmitting on frequency band 1902,as indicated by 1920. The transmitter may periodically change thefrequency band on which it operates, as described above. The number offrequency bands the transmitter on may be any number and the length oftime the transmitter transmits on any single frequency band may be thesame for all frequency bands or variable.

Also shown in FIG. 17 is an operation of the receiver during frequencyhopping. Similar to the transmitter as described above, the receiver mayoperate in three receive bands 1908, 1910, 1912. Operation of thefrequency hopping algorithm at the receiver is similar to the operationof the frequency band at the transmitter. For example, the receiver maybe receiving data on frequency band 1910. The receiver may changereceiving frequency band to frequency band 1912, as shown by 1922. Thereceiver may change which frequency band it is receiving on based on,for example, a programmed frequency hopping algorithm. After receivingdata on frequency band 1912, the receiver may again change the frequencyband on which it operates. For example, the receiver may switch fromoperating on frequency band 1912 to operating on frequency band 1910, asindicated by 1924. Thereafter, the receiver may again switch thefrequency band on which it is operating. For example, as indicated by1926, the receiver may switch from operating on frequency band 1910 tofrequency band 1908, as indicated by 1926. Thereafter, the frequencyband the receiver is operating on may change from 1908 to 1912, asindicated by 1928. The receiver may periodically change the frequencyband on which it operates, as described above. The number of frequencybands the receiver on may be any number and the length of time thereceiver receives on any single frequency band may be the same for allfrequency bands or variable.

What is material to the operations shown is that the relationshipbetween the transmit band/frequency and the receive band/frequency beknown and have been calibrated for a given pairing so that the systemcan be controllably adapted as needed during the operation of thesystem. Calibration for predetermination of parameters which can becontrolled can be done in a known automated way at the time ofmanufacture of the system. Since the system of this disclosure uses asoftware-implemented filter, the number of frequency bands and the typesof Radio Access Technologies the system may operate on is greater thanprior art systems.

Advantageously, a relationship between transmitter and receiver whereboth operate on the same frequency band can be supported within thescope of the art taught herein. Here the offset between transmit andreceive bands can be zero for either normal operation of the system orin high security or other applications. Material is that the operationfor a given transmit and receive band relationship be calibrated for thesystem including calibration for a given full duplex single frequencyoperational case.

FIG. 18 is a flowchart illustrating a method 1800 of operation of theapparatus of this disclosure, according to one aspect of thisdisclosure. The flowchart 1800 begins at steps 1802 and proceeds to1804. At step 1804, a first tone generated by, for example, a tonegenerator, is injected at the edge of the frequency band of the receivechannel. The first tone may be upconverted so that it is at whicheverfrequency band the apparatus is operating at. After completing 1804, themethod may proceed to step 1806.

At step 1806, a second tone is swept through the frequency band of thereceive channel. Like the first tone injected at 1804, the second tone1806 may be upconverted so that it is operating at the frequency bandthe apparatus is operating at. After completing 1806, the method mayproceed to 1808.

At step 1808, the apparatus may receive a first signal containinginterference. There may also be a second signal, which may be a sampleof the PA output signal. After completing 1808, the method may proceedto step 1810.

At step 1810, the apparatus may compensate the first or the secondsignal for gain and phase imbalances. For example, the apparatus may usethe I/Q compensators as described above to compensate the first or thesecond signals. In one aspect, both the first and the second signals maybe compensated using the I/Q compensators. After completing 1810, themethod may proceed to 1812.

At step 1812, the channel as seen at the receiver may be characterized.For example, the apparatus may use the swept tones at 1806 tocharacterize the channel as seen by the receiver. After completing 1812,the method may proceed to 1814.

At step 1814, the apparatus may process the first signal using aninfinite impulse response filter. The output of the infinite impulseresponse filter may be an interference-cancelling signal. Aftercompleting 1814, the method may end at 1816.

FIG. 19 is a flowchart illustrating a method 1900 of operation of thefeed-forward loop of this disclosure, according to one aspect. Theflowchart 1900 begins at step 1902 and proceeds to step 1904. At step1904, the apparatus may receive a first signal on a first path. Thisfirst signal may, for example, be a signal to be transmitted. Aftercompleting step 1904, the method may continue to step 1906.

At step 1906, the first signal may be sampled into a second path. Forexample, the first signal may be sampled using a coupler. Aftercompleting step 1906, the method may proceed to step 1908.

At step 1908, the first signal on the first path may be amplified. Theamplification, for example, may be accomplished using a power amplifier.The first signal on the first path may also be filtered using anysuitable filter, such as a band pass filter, a low pass filter, a highpass filter, or any desired combination of filters. After completing1908, the method may proceed to 1910.

At step 1910, the sampled first signal on the second path is phaseshifted in a second pathway. For example, this may be accomplished usinga linear vector modulator. The sampled first signal on the second pathmay also be filtered using any suitable filter, such as a band passfilter, a low pass filter, a high pass filter, or any desiredcombination of filters. After completing step 1910, the method mayproceed to step 1912.

At step 1912, the amplified first signal on the first path may be addedto the phase shifted, sampled first signal on the second path. The twosignals may be added, for example, using a coupler or an adder. Addingthe two signals may result in a carrier-cancelled signal on a thirdpath. The carrier-cancelled signal on the third path may have its phaseadjusted using, for example, a linear vector modulator, or its amplitudeadjusted. After completing step 1912, the method may proceed to step1914.

At step 1914, the amplified first signal on the first path may becoupled to the carrier-cancelled signal on the third path. This may beaccomplished using a coupler or an adder. After completing 1914, themethod may proceed to 1916 and end.

FIG. 20 is a flowchart illustrating a method 2000 of operation of theapparatus, according to one aspect of this disclosure. The method 2000begins at step 2002 and may continue to step 2004.

At step 2004, the apparatus may be configured to operate on a firstfrequency. After the apparatus, such as a software defined radio frontend, is configured to operate on a first frequency, the method maycontinue to step 2006.

At step 2006, the apparatus may receive a transmit signal in a firstpath. When the apparatus has received the transmit signal in the firstpath, the method may continue to step 2008.

At step 2008, the transmit signal may be amplified in the first path.The amplification may happen using, for example, a low noise amplifier.Once the transmit signal has been amplified, the method may continue tostep 2010.

At step 2010, the transmit signal may be coupled to a second path. Inthe second path, the coupled transmit signal may be phase shifted using,for example, a Linear Vector Modulator. Once the coupled transmit signalhas been phase shifted, the method may continue to step 2012.

At step 2012, the amplified transmit signal in the first path may becoupled to a third path. After the amplified transmit signal is coupled,the method may proceed to step 2014.

At step 2014, the apparatus may couple the phase-shifted transmit signalin the second path to the amplified transmit signal in the third path.This coupling may result in a carrier-cancelled signal in a fourth pathin the apparatus. After the signals in the second and third path havebeen coupled, the method may continue to step 2016.

At step 2016, the apparatus may phase shift the carrier-cancelled signalin the fourth path using, for example, a Linear Vector Modulator. Oncethe carrier-cancelled signal in the fourth path is phase shifted, themethod may continue to step 2018.

At step 2018, the apparatus may couple the phase shifted,carrier-cancelled signal in the fourth path with the amplified transmitsignal in the first path. After the phase shifted, carrier-cancelledsignal in the fourth path and the amplified transmit signal in the firspath are coupled, the method may continue to step 2020.

At step 2020, the apparatus may be reconfigured to operate on a secondfrequency. After the apparatus has been reconfigured to operate on thesecond frequency, the method may continue to step 2022 and end.

FIG. 21 is a flowchart illustrating a method 2100 of operation of theapparatus, according to one aspect of this disclosure. The method 2100begins at step 2102 and may proceed to step 2104.

At step 2104, the apparatus may upconvert a first tone a first frequencyand a second tone to a second frequency. The first and secondfrequencies may be pre-calibrated to operate in a plurality of frequencybands. The first frequency may overlay the frequency band of the receivechannel. Additionally, the apparatus may upconvert a third tone to afifth frequency. After completing step 2104, the method may proceed tostep 2106.

At step 2106, the apparatus injected the upconverted first tone into areceive channel. After completing step 2106, the method may proceed tostep 2108.

At step 2108, the apparatus may sweep the upconverted second tonethrough the frequency band of the receive channel. The number of stepswithin the sweep may be configurable by the controller. Also, thefrequency band of each step may be configurable by the controller. Aftercompleting step 2108, the method may proceed to step 2110.

At step 2110, the apparatus may use the swept upconverted second tone tocharacterize the receive channel. After completing step 2110, the methodmay proceed to step 2112.

At step 2112, the apparatus may process the transmit signal using an IIRfilter. The coefficients of the IIR filter may be generated using thecharacterized receive channel. Additionally, the IIR filter coefficientsmay be generated in conjunction with a least-squares estimator circuit.The IIR filter may output an interference-cancelling signal.Additionally, the apparatus may couple the interference-cancellingsignal to a received signal containing interference. After completingstep 2112, the method may proceed to step 2114.

At step 2114, the apparatus may upconvert the first tone to a thirdfrequency and a second tone a fourth frequency. The third and fourthfrequencies may be in a different frequency band from the first andsecond frequencies. The first, second, third, fourth, and fifthfrequencies may be determined based on the radio access technology theapparatus may be operating on. After completing step 2114, the methodmay end at step 2116.

The present description is for illustrative purposes only, and shouldnot be construed to narrow the breadth of the present disclosure in anyway. Thus, those skilled in the art will appreciate that variousmodifications might be made to the presently disclosed embodimentswithout departing from the full and fair scope and spirit of the presentdisclosure. Other aspects, features and advantages will be apparent uponan examination of the attached drawings and appended claims.

1. A self-interference cancellation system, comprising: a transmitterpower amplifier configured to amplify a transmit signal; a firstcirculator directly coupled to the transmitter power amplifier toreceive the transmit signal, the first circulator being coupled to atransmission-reception path of the self-interference cancellationsystem; an antenna at an end of the transmission-reception path, whereinthe antenna is configured to transmit the transmit signal and receive areceive signal; a receiver amplifier coupled to a reception path of theself-interference cancellation system; a second circulator coupled tothe first circulator; and a filter coupled to the second circulator inthe reception path.
 2. The system of claim 1, further comprising: aphase shifter located between the filter and the second circulator. 3.The system of claim 1, further comprising: an electrical load, whereinthe filter is located between the second circulator and the electricalload.
 4. The system of claim 2, further comprising: an electricalground, wherein the filter is located between the phase shifter and theelectrical ground.
 5. The system of claim 1, wherein the secondcirculator is located between the antenna and the first circulator alongthe transmission-reception path.
 6. The system of claim 1, wherein thesecond circulator is located between the first circulator and thereceiver amplifier along the reception path.
 7. The system of claim 4,further comprising: an attenuator coupled to the filter.
 8. The systemof claim 1, wherein the filter is a transmitter bandpass filter or areceiver notch filter.
 9. The system of claim 3, wherein the electricalload includes a resistor.
 10. The system of claim 1, wherein the systemis a wireless communication base station or a mobile wireless device.11. The system of claim 1, wherein a phase shifter shifts a phase of areflected signal from the antenna a number of degrees to cancel aportion of the transmit signal that entered the receive path towards areceiver from the first circulator.
 12. A method of performingself-interference cancellation comprising the steps of: generating atransmit signal along a transmit path of a transceiver; sending thetransmit signal to a power amplifier and then directly through acirculator to substantially isolate the transmit signal from a receiver,wherein at least a portion of the transmit signal enters a receive pathtowards the receiver; generating a reflected signal from the antenna,wherein the reflected signal from the antenna is at substantially lesspower than an incident power to the antenna, wherein the reflectedsignal includes a transmitter carrier signal and a transmitter noise;routing a received signal from the antenna to the receiver; routing thereflected signal through a filter, wherein a reflected transmittercarrier signal is passed through the filter to a load; and combining thereflected transmitter noise with the received signal in the receive pathto cancel the portion of the transmit signal that entered the receivepath towards the receiver from the circulator.
 13. The method of claim12, wherein the circulator is a first circular, and wherein the methodfurther comprises a second circulator coupled to the first circulator.14. The method of claim 13, wherein the second circulator is locatedbetween the antenna and the first circulator along atransmission-reception path.
 15. The method of claim 14, wherein thesecond circulator is located between the first circulator and a receiveramplifier along a reception path.
 16. The method of claim 13, whereinthe phase shifter is coupled to a variable attenuator, where thevariable attenuator adjusts an amplitude of the reflected signal. 17.The method of claim 16, wherein the filter is a transmitter bandpassfilter to remove the transmitter carrier signal in the receive path. 18.The method of claim 16, wherein the filter is a receiver notch filter toreflect the received signal in the receive path.
 19. The method of claim16, further comprising: routing the reflected signal through a phaseshifter in the receive path, wherein the phase shifter shifts a phase ofthe reflected signal from the antenna a number of degrees to cancel theportion of the transmit signal that entered the receive path towards thereceiver from the circulator.
 20. The method of claim 16, wherein thephase shifter and the variable attenuator are adjusted by a feedback andcontrol algorithm applied by a controller.